Motorcycle Battery Charger

This 3A charger was originally designed to work with small batteries like those used in motorcycles. In principle it can be used to charge car batteries also but will take a lot longer.
The charger below charges a battery with a constant current to 14.1 volt. When this level is reached, the current charge drops automatically to a safer level (13.6V) and keeps charging at this slower rate untill the LED lights up indicating a fully charged battery. This project looks very much alike with the Gel cell II charger elsewhere posted in the 'Circuits' section. The difference is the IC, namely a LM1458 instead of a LM301A. Nice job Jan!



Description:

The LM350 is an adjustable voltage regulator and keeps the voltage between points C and B at 1.25 Volt. By adding a 1K resistor between point B and gnd (-) you can, as it were, lift up the output voltage. To accurately control the output voltage we add to this resistor, in series, a 2K adjustable 10-turn potentiometer. As soon as a battery is connected a current flow occurs, controlled by the right halve of the LM1458. The current through the 0.1 ohm resistor causes a voltage drop. This drop is compared with the voltage on the walker of 100-ohm pot. The moment this drop is greater than the one adjusted with the potmeter will cause the output of the LM1458 IC to go low and a small current starts to flow thru the diode and this in effect will reduce the current through the series resistors 1K + 2Kpot. The current is hereby stabilized.

The point between C and B is devided by three resistors; 2.2 ohm, 100 ohm pot, and the 150 ohm. 2.2 ohm and the 100 ohm potmeter are connected to the non-inverting input (+) of the LM1458 IC. The inverting input (-) is connected to the 0.1 ohm wire-wound resistor in series with the output. As long as the voltage drop, caused by the current-flow over this resistor is greater than the voltage drop over the 2.2 ohm resistor the output of the LM1458 will stay high and in turn block the BC558 transistor. But as soon as the charge current falls below a specific value the 1458 will go low and turn on this transistor which wich activate the LED. At the same time a small current will flow thru the 'Rx' resistor, which will cause that the output voltage of the charger switches to 13.6 Volt. This is a very safe output voltage, and does not cause overcharging to the battery and remains fully charged (trickle).

Rx should be an experimental value determined below; a mathematically calculation is possible but the exact value is determined by the tolerances of your specific components.

The voltage regulator LM350 has to dissipate a lot of energy so make sure to mount it on a large cooling fin. (e.i. 3.3°C/Watt) Diode 1N4001 over the input/output is necessary to prevent damage to the regulator in case the input voltage gets interrupted.

The LM350 can be substituted with a NTE970, and the BC558B with a NTE159 if you wish.
The adjustments for this charger are really simple and the only thing needed is digital multimeter. The LM1458 should NOT be in the socket while doing the first adjustment. When no battery is connected there is no current flow thru the 0.1 ohm resistor and therefore pulling the output low. So no IC yet in the socket. Do NOT connect a battery also. I know that is obvious to most of us, but some people... :-)

Okay, here we go:

1. Connect the multimeter (set for Volt DC) to the '+' and '-' battery output and adjust with the 2k trimpot the output voltage to 14.1 Volt.
2. Switch the power off. Discharge the capacitors (short them out with a piece of wire).
3. Now insert the LM1458 IC carefully (check no pins are bend underneath the chip).
4. Switch the power back on and make the resistor marked Rx such a value that the output voltage reads 13.6 volt exactly.
5. Switch the multimeter to 'Amp-dc'. Turn the 100-ohm trimpot all the way CCW. Connect the 'to-be-charged-battery' (e.i. NOT a fully charged battery) and turn back the trimpot untill the current load is 0.1 X the battery capacity (max 3A). Example: A 16Amp battery adjusting to 1.6A. If you don't have an Amp meter on your multimeter you can use the 2-volt setting on your meter and connect it over the 0.1 ohm resistor. The current is volt devided by 0.1, so for 3A the meter should read 0.3 volt.

That's it. To get the Rx value you could also use a trimpot until you get the 13.6volt and then read the ohm's value of the trimpot and replace with a resistor. In my opinion this resistor should be a metalfilm type at 1 or 2% tolerance.

The Technical bits:

For those of you interested in how the value of essential components was calculated, read on. You may be able to design your own charger for use with a different current or voltage (like 6-volt).
Calculations origin from the voltage between points C and B of the LM350 regulator. When a resistor is connected between these two points, enough current starts to flow that the voltage over this resistor measures 1.25 volt. In our case, the resistor total is 2.2 + 100 + 150 =252.2 ohm. Because we deal with very small currents the calculations are performed in milliamps and the calculations of resistance in Kilo-Ohms. Thus, the current thru this resistor is 1.25 / 0.2522 = 4.9564 mA. The same current also flows thru the 1K & 2K series resistors. We want the output voltage to be 14.1 volt, meaning the voltage drop over these series resistors must be 14.1 - 1.25 = 12.85 Volt.

The total resistance value thus must be 12.85 / 4.9564 = 2.5926 Ohms. To enable us to adjust it to this value, one of the resistors is chosen as a 10-turn trimpot (trimmer potentiometer). Together with the 1K in series (making it a total of 3K)we can adjust it to this correct value.

The Rx value is calculated this way; In this scenario we like to have a output voltage of 13.6 volt, in other words, the voltage on the connection point between the 1K/2Kpot should be 13.6 - 1.25 = 12.35 volt. This means that the current thru the 'voltage-divider' will be 12.35 / 2.5926 = 4.7635 mA and the leftover current should be 4.9564 - 4.7635 = 0.1929 mA thru Rx and also cause a voltage drop of 12.35 - 2.78 = 9.57 volt. Measuring this calculated value at the base of the BC558 transistor was 2.78 volt after the output of the LM1458 had become low. With the current of 0.1929 mA the result has become9.47 / 0.1929 = 49.611 Kilo-Ohm. A resistor of 47K would come close enough. Ofcourse you could also use a 50K trimpot to adjust the value even more accurately. The 1K5 (1500 Ohm) resistor in series with the LED is to limit the current thru the LED below 20 mA.

The only thing left is to calculate the value of the series resistor which determines the switch-over from charge to float condition. This occurs when the voltage drop over the 0.1 ohm (wire-wound) resistor at the positive leg smaller is than over the 2.2 ohm resistor. This value is 2.2 x 4.9564 = 10.9 mV. The resistance is 0.1 ohm, to get a voltage drop over this resistor of 10.9 mV is the current 10.9 x 0.1 = 109 mA. The second this charge current becomes lower then 109 mA, the LM1458 triggers over to the float condition.
The adjustment with the 100-ohm trimpot determines the maximum charge current. The voltage on the walker of this trimpot varies between 10.9 mV - 506.54 mV. The current is this way made adjustable between 0.1A - 5A, but we should not go that far because the LM350K can not handle anything over 3Amp. If we chose a trimpot with a value of 50 ohm, then on the other hand the 3A can not be obtained. So, careful adjustment is the remedy. Take your time!

With this information it is a simple task to calculate the dissipation values of the resistors. In other words, the product of the resistance multiplied with the current in square (I2xR).

The only resistor which gets it difficult is the 0.1 ohm, but then again, not by much 3 x 3 x 0.1 = 0.9 Watt.
Rest us to calculate the power. For that we have add a couple of voltages. We have the input voltage of 14.1, the voltage drop over the resistor, 0.1 x 3 = 0.33 volt, and 3 volt minimum over the LM1458 for proper function, total 17.43 volt. The transformer provides 18V (effective). With ideal rectifying this should total 18 x 1.41 = 25.38 volt. There are however losses via the diodes and bridge rectifier so there is about 23.88 volt remaining. Not much tolerance to play with, on the other hand, too much causes energy loss in the form of heat anyway.
The voltage drop over the buffer capacitor may not be lower than 17.43 volt, meaning, the ripple voltage may reach about 23.88 - 17.43 = 6.45 volt. By double-fase rectifying is the ripple voltage equal to I/(2xfxC) whereby I is the discharge current, f is the supply frequentie and C is capacity of the buffer capacitor in Farad. Exchanging places this would give C = 3/(2x50x6.45) = 0.004651 Farad, or 4651 uF. A standard value of 4700 uF with a minimum voltage value of about 35-40 Volt. The other capacitor is not very critical and is only there to kill small voltage spikes which could influence the operation of this charger otherwise.

The bridge rectifier gets a good workout also and it is therefore recommended to chose NOT a too light a unit. A 5A rectifier is often too small, better to take a 8 or 10A type. These are readily available everywhere.

Last but not least, the transformer. The buffer capacitor has approximately 25 volt accros. The current is 3A. This calculates to a power of 25 x 3 = 75 watt. This transformer has its own problems with powerloss (naturely occuring) and so a unit of about 80 watt is acceptable.
Never attempt to charge a 6 volt battery with a 12 volt charger; you are asking for trouble. Good luck all!

Foam Cutting Power Supply

After seeing other modelers building their model wings from plastic foam, I decided that I wanted to do the same. Building your wings from foam covered with 1/16 in. balsa can produce a strong and light wing that could be difficult to duplicate with the standard balsa rib construction, especially if the wing had a duel tapered, symmetrical airfoil. The standard way to cut foam is with the Hot Wire technique, using steel or nichrome wire through which an electrical current flows to heat the wire.


However, the methods that many use to get the wire hot leaves something to be desired. The most common method I saw used was to connect a 12volt battery charger to 4 or 5 feet of nichrome wire which was tied to some kind of a bow. Using the variable charging rate, you could control (to a limited degree) the temperature of the wire and thus the speed of the cut. But if you cannot accurately control the heat, you'll get many poor cuts. Some have connected a series of light bulbs in line with the wall service of 115 volts AC.

It works, but WOW, is it ever dangerous! Terrible shock hazard! I've even seen some connect the nichrome wire across a 12 volt car battery, also very dangerous. Over the years there have been several schematics listed in the model magazine for building a hot wire foam cutter power supply. All of them worked, I'm sure. Some were very simple, but left little heat control, and others were complex and expensive. Heat control is the secret for making good foam cuts. Also a good transformer is important for removing the electrical shock hazard that threatens the modeler in his shop. A good current limiting feature also makes the device safe from high current burns, which some auto mechanics have suffered when working with large 12 volt batteries.


The following circuit is a simplification of several older designs. This design uses readily available parts, is easy to build, has total temperature control for both a long bow (48") and a short bow (24"), and has served me well for the last 15 years. Many of the planes that I fly are my own design and I build most of them with foam wings, foam turtle decks, foam stabs, etc, usually with dual tapered, symmetrical designs. The short bow is valuable for sculpting foam pieces into various shapes, as it can be held in one hand and the foam sample in the other.



The first step in building one of these foam cutters is to take the Bill of Materials to your local Radio Shack and search for the parts. I picked this source because of shopping convenience and the total cost is a little above $30. Also get a small copper clad circuit board (CB), about 3 by 4 inches or larger in size. If you chose not to make the circuit board, you can solder the parts together using electrical stand-offs. The first order of business is to mount the switches, the potentiometer, the Red and Black electrical posts (#274-662), and the red indicator light on the front panel of the component box according to the picture and illustration. Next mount the transformer (#273-1512), fuse holder (#270-364), and electrical cord (#278-1255) in the box as shown in photo. Put some rubber feet on the bottom of the box (also from Radio Shack) so that it won't scratch your wife's end table when you take it to show her what a great craftsman you are.


The circuit is a simple AC Triac voltage control circuit similar to the ones used to control house lamps. The transformer provides the electrical isolation the makes this item safe to operate. The voltage at the bow will tingle a little, but will not harm the operator. The OFF/ON switch is a simple s.p.s.t. switch (#275-651). The "Long/Short" Bow selector switch is the same part number. Across the primary side of the transformer is mounted an indicator "ON" lamp (#272-712) which will light up when the unit is turned on. The temperature control is through the 5k ohm potentiometer R2 (#271-1714). The Triac gate current is controlled by R3, a 470 ohm, 1 watt resistor. This resistor is not part of radio Shack's inventory, therefore it may be required to solder two 1k ohm, 10 watt resistors in parallel. The capacitor, C1, is a 0.22microF disk (#272-1070). The 5 ohm, 20 watt resistor R1 is made of two 10 ohm, 10 watt resistors in parallel. They are large ceramic resistors mounted side by side. These resistors drop the voltage when the short 24" bow is being used. These resistors will get hot, don't touch!

Enclosed in this article is a actual size drawing of the circuit board (CB, 2.5" x 4"). Cut out this drawing and use it as a template, and paste it on the side opposite of the copper on the CB (circuit board) with some rubber cement. Next use a center punch to mark the center of each hole. Then drill the holes with the CB held tightly to some wood backing, making the four corner holes a 1/8" in dia and all the rest about 1/16" in dia. These smaller holes will be where you solder the components and wires. The larger holes are for the mounting bolts to hold the CB to the case. Cut the CB to the exact size as shown on the template (2.5" x 4"). Then, print out the copper side drawing and paste it on the copper side of the board. Use a sharp X-acto knife to remove thin strips of copper as shown. This will isolate the copper soldering pads from one another. Remove the paper. Insert the components in the CB on the side opposite the copper. Where the component leads stick out on the copper side, solder the component leads to the board being careful not to allow solder to bridge the cut lines in the copper. Cut off any excessive lead after soldering it. Bolt the Heat Sink on to the Triac with the fins pointing out. The Triac should be mounted in a vertical position, perpendicular to the CB. Next mount the CB in the box with 6-32 x 1" bolts and stand-offs. Finish soldering the connecting wires to the board before tightening the bolts. Drill 3 or 4 vent holes (1/4" dia) in the top of the box in the area above the Triac heat sink.

Plug the nichrome wire bow leads into the dual plug speaker connectors. It is best to trim the wire insulation on the wires back about 1/2 in. then tin the wire ends. After the wires are plugged in, turn the unit on and with the temperature control at half point and the heat switch set up to long. The wire should get hot to the touch almost immediately. If it doesn't, then examine the construction on the circuit board and wiring, and fix any errors found. After the unit is finished, bolt the top on and your and you're done.

When you use the foam cutter, be sure that the Bow switch is in the correct position. The switch must be in the "Short" position (down) if the 24" bow is used. Otherwise you may blow the fuse. Leave the unit in the "Long" bow position at all times unless you are using the short bow and you should have no problems. Before you turn the Foam Cutter on, turn the temperature (TEMP) control fully counter-clockwise, to minimum temperature. Turn the unit on with a bow plugged in and increase the temperature by turning the TEMP knob clockwise. The temperature of the wire increases almost immediately. With a piece of foam, test for the foam cutting temperature. Reduce the TEMP control until the cut is smooth with little foam evaporation around the wire. Remember, the smoothest cuts are made slowly. Spend some time practicing until your cuts are smooth. You will never go back to balsa ribs!

Lead Acid Battery Charger

Parts List:

C1 = 100uF/63V
C2 = 10uF/63V
D1 = 1N5401/NTE5801
D2 = LED (Red, 5mm)
Q1 = NTE374/BD140
Q2 = NTE123AP/BC547
R1 = 120 Ohm
R2 = 82 Ohm
R3 = 10K
R4 = 33K
R5 = 22K
P1 = 2K2
U1 = LM350 (On large coolrib!)


How it works:

Except for use as a normal Batter Charger, this circuit is perfect to 'constant-charge' a 12-Volt Lead-Acid Battery, like the one in your flight box, and keep it in optimum charged condition. This circuit is not recommended for GEL-TYPE batteries since it draws to much current.

The above circuit is a precision voltage source, and contains a temperature sensor with a negative temperature coëficient. Meaning, whenever the surrounding or battery temperature increases the voltage will automatically decrease. Temperature coëficient for this circuit is -8mV per °Celcius. A normal transistor (Q1) is used as a temperature sensor.

This Battery Charger is centered around the LM350 integrated, 3-amp, adjustable stabilizer IC. Output voltage can be adjusted with P1 between 13.5 and 14.5 volt. T2 was added to prevent battery discharge via R1 if no power present. P1 can adjust the output voltage between 13.5 and 14.5 volts. R4's value can be adjusted to accommodate a bit larger or smaller window. D1 is a large power-diode, 100V PRV @ 3 amp. Bigger is best but I don't recommend going smaller.

The LM350's 'adjust' pin will try to keep the voltage drop between its pin and the output pin at a constant value of 1.25V. So there is a constant current flow through R1. Q1 act here as a temperature sensor with the help of components P1/R3/R4 who more or less control the base of Q1. Since the emitter/base connection of Q1, just like any other semiconductor, contains a temperature coëficient of -2mV/°C, the output voltage will also show a negative temperature coëficient. That one is only a factor of 4 larger, because of the variation of the emitter/basis of Q1 multiplied by the division factor of P1/R3/R4. Which results in approximately -8mV/°C. To prevent that sensor Q1 is warmed up by its own current draw, I recommend adding a cooling rib of sorts.
(If you wish to compensate for the battery-temperature itself, then Q1 should be mounted as close on the battery as possible) The red led (D2) indicates the presence of input power.

Depending on what type of transistor you use for Q1, the pads on the circuit board may not fit exactly (in case of the BD140).

Caution: Adjust the voltage of capacitor C1 according to the input voltage. Example, if your input voltage will be 24 volt, your C1 should be able to carry at least 50V.

IC Controlled Emergency Light with Charger

The circuit shown here is that of the IC controlled emergency light. Its main features are: automatic switching-on of the light on mains failure and battery charger with overcharge protection. When mains is absent, relay RL2 is in deenergised state, feeding battery supply to inverter section via its N/C contacts and switch S1. The inverter section comprises IC2 (NE555) which is used in stable mode to produce sharp pulses at the rate of 50 Hz for driving the MOSFETs. The output of IC3 is fed to gate of MOSFET (T4) directly while it is applied to MOSFET (T3) gate after inversion by transistor T2. Thus the power amplifier built around MOSFETs T3 and T4 functions in push-pull mode.



The output across secondary of transformer X2 can easily drive a 230-volt, 20-watt fluorescent tube. In case light is not required to be on during mains failure, simply flip switch S1 to off position.

Battery overcharge preventer circuit is built around IC1 (LM308). Its non inverting pin is held at a reference voltage of approximately 6.9 volts which is obtained using diode D5 (1N4148) and 6.2-volt zener D6. The inverting pin of IC1 is connected to the positive terminal of battery. Thus when mains supply is present, IC1 comparator output is high, unless battery voltage exceeds 6.9 volts. So transistor T1 is normally forward biased, which energises relay RL1. In this state the battery remains on charge via N/O contacts of relay RL1 and current limiting resistor R2. When battery voltage exceeds 6.9 volts (overcharged condition), IC1 output goes low and relay RL1 gets deenergised, and thus stops further charging of battery. MOSFETs T3 and T4 may be mounted on suitable heat sinks.

New Vacuum Tube Amplifier

Features

• Output: 4-6550's in triode-mode class AB2 push-pull parallel. About 80 watts RMS per channel.
• No global negative feedback. Several local loops with limited negative feedback.
• Ultra-wide bandwidth Plitron toroidal output transformer.
• Servo to maintain precise dc-balance in the output circuit.
• MOSFET-regulated power supplies. Relative rather than absolute voltage reference.
• Designed using extensive PSpice computer simulation.
• Constructed as a pair of monoblocks.

Schematic diagram


Notes on the schematic diagram

• The power supply has been simplified-- Power transformers and rectifiers have been omitted and some parts have been omitted from the MOSFET voltage regulator circuits: 1N5242 zener diodes between the source and gate and 10k resistors in series with the gate. These parts serve as protection in case of accidental short circuits, but don't affect the operating point. The full power supply schematic is shown below.
• 6SN7's are used instead of 12AU7's for the driver tubes. They have the same plate characteristics, but they have higher maximum plate voltage (450 vs. 330 V) and greater plate dissipation (3.75 vs. 2.75 W per section). Think of the octal-based 6SN7 as a 12AU7 on steroids.
• The NODESET blocks (lower right) initialize dc levels in the bias servo so the simulation runs properly. They don't exist as physical entities.
• All resistors are 1/2 watt unless noted, and with the following exceptions. R3P, R4P and R5C through R8C (68K) are 2W. R9C through R12C (20 ohm) are 1W: I used 2-10 ohm resistors in series to make them. R9S through R12S (200 ohms) are 2W.
• Capacitor voltage ratings: It never hurts to go over the minimum, though the capacitors will be larger and may cost more. I often use caps with higher voltage ratings because I have them on hand or found them at a good price in an electronics surplus shop. Here are some minimum ratings: C1G: 100V. C2G: 400V (600V would be better; necessary without the time delay); C3G and C4G: 400V (600V would be better); C3M and C4M: 400V; C5G through C8G: 600V; CBS2, CBS4, CBS6 and CBX1 through CBX3: 100V. The voltage ratings of many of the power supply capacitors are shown on the circuit board wiring diagrams, below.
• Some of the feedback connections may be a little hard to trace. ORN goes between the 20 ohm output transformer secondary feedback winding and R3F near TU3. Similarly, VLT goes to R4F near TU4. BLU and BLK on the output transformer secondary speaker winding go to the output tube cathode circuits.
• RLS (5 ohms) is a simulated speaker load.
• CRF: All three nodes are connected together.
• BRN and VIO are not used. They are the ultra-linear taps. In the original version of TENA there was a switch to select output tube screen grid connections between BRN and VIO (UL mode) and GRN and YEL (triode mode).

Input stage/phase inverter

Input stage TU1 is a simple voltage gain stage with local negative feedback, derived from the R1B, R1C voltage divider. It is capacitively coupled to split load phase inverter TU2. The capacitor has an unusually low value-- 0.01 µF-- because TU2 has an exceptionally high input impedance-- several Megohms. The advantage of capacitive coupling is that it allows the voltage level in TU2 to be set for maximum output and it allows the ac current in TU2 to be precisely equal to, but 180 degrees out of phase with, the current in TU1. The net ac current drawn by these two tubes from V+420 is therefore zero. This is an effective way of isolating the audio signal from the power supply, which doesn't need to supply ac current. In conventional designs ac signal often has to flow through electrolytic capacitors, which are grungy leaky devices with memory-- harmful to audio quality. I designed TENA to draw zero net ac current from all power supply outputs (easy to do in a push-pull design), at least up to the power level where one of the output tube pairs starts cutting off.

Toroidal output transformer

We chose the Plitron toroidal transformer because of its exceptional bandwidth: -3 dB at over 200 kHz, the result of high primary inductance (the good stuff) and low leakage inductance (the bad stuff-- kind of like HDL and LDL cholesterol)-- much better than can be achieved with a conventional EI transformer. High bandwidth is important because output transformers have an intrinsic second order rolloff, which can make them unstable in the presence of negative feedback unless careful phase compensation is applied (see Feedback and Fidelity). Phase compensation reduces the bandwidth, which is not a problem with the Plitron toroidal transformers. But this bandwidth comes at a price-- toroidal transformers are much less tolerant of dc-imbalance than EI transformers; they may saturate at dc imbalances as low as 8 mA. (I don't know the exact number; I never simulated it.) You would have to set the bias of each tube individually, and then you'd have to worry about how the tubes age. So I designed a bias servo circuit to maintain perfect dc-balance under all circumstances except outright tube failure.

The Plitron PAT 4006CFB 100 Watt toroidal output transformer is not currently listed on Plitron's website, but I've heard (June 2003) that it is available. Contact Norman Woo. The closest models are the 4006, which lacks the special feedback winding, and the 2100-CFB which has a higher primary impedance. The minimum feedback version of TENA (below) works with the 4006.

Bias servo and adjustment

The time-averaged (low pass filtered) dc current of an output tube operating in class AB fixed bias is relatively constant at low power levels but increases at high power levels. For this reason a fixed voltage cannot be used as a reference for biasing the output tubes. One tube (TU9, driven by TU5) operates at fixed bias, and its low pass filtered cathode voltage (CRF) is used as the reference for biasing the other tubes.

The bias servo is illustrated in the lower left of the schematic. It uses the LM324 quad op amp-- cheap but perfectly adequate. Inputs U1A, U1B and U1C of the LM324 compare cathode voltages 10C, 11C, and 12C with reference voltage CRF, which is the voltage on cathode 9C low pass filtered with RBS2 = 33k and CBS1 = 10µF ( located near U1B on the schematic). The LM324 outputs control the P-channel MOSFETs, each of which controls a voltage divider between VBB (-90V) and VOP (+12.5V) to deliver the appropriate bias voltage to the driver grid circuits (BIAS_6, BIAS_7, and BIAS_8). This measures between -45 and -50V in my amplifiers, which operate at 60 mA plate current. Audio purists please note: the servo operates at extremely low frequencies; the op amp and MOSFETs are well outside the audio signal path.

A single potentiometer, RB5 (in the VBB supply, bottom center), controls the bias current directly in TU9, and all the other tubes indirectly through the servo. Bias current may be measured across any of the 20 ohm resistors R9C-R12C as E/20. They should all be the same if the servo is working properly. 1 to 1.2 volts is a good nominal value, corresponding to 50 to 60 mA per tube (70 mA was used in the Dynaco Mark III). Increasing the current increases power consumption and reduces tube life and output power, but moves you closer to Class A (where both tubes always conduct).

Class AB2 output stage and drivers

Class AB2 differs from the more common class AB1 in that the output stage grid is driven positive-- it draws grid current-- at high power levels. Class AB2 has no advantage for output tubes operating in pentode mode and little advantage for ultra-linear mode. But it results in a huge power boost for output tubes operating in triode mode. You can get almost as much power out of class AB2 triodes as you can out of class AB1 pentodes.

If you try to do operate in class AB2 with conventional capacitive coupling, the coupling capacitor starts charging as soon as grid current is drawn. This drives the grid negative-- toward cutoff, and it recovers with the RC time constant of the coupling capacitor and grid resistor. To operate successfully in class AB2, the output stage must be either transformer or direct coupled. I chose direct coupling because interstage transformers are expensive and have limited bandwidth.

The direct coupled drivers are the source of much of TENA's complexity. Because the quiescent grid voltage of each output tube must be set individually to control its quiescent (dc) current, one driver tube (TU5-TU8) is required for each output tube (TU9-TU12). Cathode followers (CF's) were chosen because they have low output impedance and can source the needed output tube grid current. The cathodes have to be somewhere near -50V to properly bias the output tubes. This means the CF must be driven by voltages outside the range of conventional power supplies, hence the need for VDR- and VDR+: the price of perfection. In reviewing the design I find that the driver tubes may be operating a little too conservatively-- dissipating only 0.78 W (of a 6SN7 maximum of 3.75 W). I've discussed driver dissipation under PSpice output, below. I may increase VDR+ from 205 to around 250 V by increasing RD1 from 470k to 680k. This would reduce the power dissipation in MOSFET MD1.

Output tube grid stop resistors R9G-R12G play an important role in TENA's soft clipping. When power levels become high enough level for grid current to be drawn, a voltage drop across these resistors gradually limits the plate current. Soft clipping consists of low order harmonics which have much less adverse effect on sound quality that the high order harmonics characteristic of abrupt clipping. But total harmonic distortion for soft clipping amplifiers tends to be higher. Yes, lower harmonic distortion doesn't mean better sound. See "The great harmonic distortion scam" in Feedback and Fidelity. TENA oscillated when the grid stop resistors were removed. This was the only performance feature PSpice didn't catch. The reason is that the output transformer model is somewhat simplified-- it's extremely difficult to model its distributed capacitance.

Power supplies

The time delay circuit (U3 (the 555B chip), Q1, Relay_SPDT_nb, RT1, CT1, CT2, RT3, D1, RV1, and RT4) has apparently never been implemented. RT4 should be replaced by a straight wire; VBIN is connected directly to NTC (negative temperature coefficient; 50 ohms cold; Mouser527-3504-50) thermistor RV10.

The precise values of most of the capacitors in the power supply, particularly CV1, CV2, CB1, CB2, CD1 and CD2, are not critical. In many cases they were determined by parts availability. If the values are 2 uF or under they are film capacitors. If they are over 2 uF they are electrolytics.

Depending how you count there are two (power transformers), four (rectifier circuits) or six (voltage levels). All use fast recovery rectifier diodes. All except VDR- are taken from the mighty Plitron 854710 toroidal power transformer, which I can't seem to find in their catalog. Toroidal power transformers perform well, but they have less of an advantage than toroidal output transformers-- you don't need wide bandwidth for 60 Hz. The CL80 inrush current limiter limits turn-on current in the tube filaments.

Universal Input Linear Fluorescent Ballast

Features

• Drives one 35 W TL5 Lamp
• Input Voltage: 80 VAC to 260 VAC
• High Power Factor/Low THD
• High Frequency Operation
• Lamp Filament Preheating
• Lamp Fault Protection with Auto-Restart
• Low AC Line Protection
• End of Lamp Life Shutdown
• IRS2166D(S)PbF HVIC Ballast Controller

The Board is a high efficiency, high power factor, fixed output electronic ballast designed for driving rapid start fluorescent lamp types. The design contains an EMI filter, active power factor correction and a ballast control circuit using the IRS2166D(S)PbF Ballast Control IC1.

The Board consists of an EMI filter, an active power factor correction section, a ballast control section and a resonant lamp output stage. The active power factor correction section is a boost converter operating in critical conduction mode, free-running frequency mode. The ballast control section provides frequency modulation control of a traditional RCL lamp resonant output circuit and is easily adaptable to a wide variety of lamp types. The ballast control section also provides the necessary circuitry to perform lamp fault detection, shutdown and auto-restart.

This board is designed for single TL5/35W Lamp, voltage mode heating (JV1 and JV2 mounted, JC1 and JC2 not mounted). TL5 lamps are becoming more popular due to their lower profile and higher lumen/ watt output. These lamps, however, can be more difficult to control due to their higher ignition and running voltages. A typical ballast output stage using current-mode filament heating (filament placed inside L-C tank) will result in excessive filament current during running. The output stage has therefore been configured for voltage-mode filament heating using secondary windings off of the resonant inductor LRES. The lamp has been placed outside the under-damped resonant circuit loop, which consist of LRES and CRES. The filament heating during preheat can be adjusted with the capacitors CH1 and CH2. The result is a more flexible ballast output stage necessary for fulfilling the lamp requirements. The DC blocking capacitor, CDC, is also placed outside the under-damped resonant circuit loop such that it does not influence the natural resonance frequency of LRES and CRES. The snubber capacitor, CSNUB, serves as charge pump for supplying the IRS2166D.

The IRS2166D Ballast Control IC is used to program the ballast operating points and protect the ballast against conditions such as lamp strike failures, low DC bus, thermal overload or lamp failure during normal operations. It is also used to regulate the DC bus and for power factor control allowing high power factor and low harmonic distortion.

Switch Mode Power Supply 100W - 16 at 7A

Generally, Schottky diodes are traditional devices use in passive rectification in order to have low conduction loss in secondary side for switching power supplies. The proliferations of synchronous rectification (SR) idea - which is mostly use in buck-derive topologies - have reached the domain of flyback application in recent years. The use of low-voltage-low-Rdson mosfet has become so attractive to replace the Schottky rectifiers in high current applications because it offers several system advantages such as dramatic decrease in conduction loss and better thermal management of the whole system by reducing the cost investment in heat sink and PCB space.



A number of techniques in the implementation of SR in flyback converters are continuously growing from a simple self-driven (secondary winding voltage detection) to a more complex solution using “current transformer sensing” or combinations of both to improve the existing technology. The idea has become quite complicated though and additional discrete devices have made the cost and part counts issue even worse. Moreover, the issue of reverse current conduction (-due to the delay in sensing the sharp drop of secondary current during turn-off phase of the SR) still lingers on in different input line/ output load conditions. The use of a simple fast-rate-direct-sensing of voltage drop across the mosfet (Vsd) using integrated solution has pave the way for a much simpler and effective means of controlling the SR mosfets as well as alleviating the reverse current and multiple-pulse gate turn-ON issues.

The board is a universal-input flyback converter with single DC output capable of delivering continuous 100W (@ +16V x 6.25A) during active rectification mode. This board is primarily designed to study synchronous rectification using IR1166 in low-side configuration to take advantage of simpler derivation of Vcc supply from converter’s output. It is equipped with necessary jumpers to ease exploring the conduction behavior of synchronous rectifiers SRs in quasi-resonant mode, so discussion would be confined to variable frequency switching in Critical Conduction Mode.

It features the fast Vsd sensing of the IR1166 Smart Rectifier Control IC with gate output drive capability of 1.5Apk. It drives 2 pcs. of SRs in parallel (100V N-ch mosfet IRF7853 in SO-8 package with very low Rdson in its class : 18 mΩ max). This had greatly simplified the overall mechanical design for not having those bulky and heavy heat sinks normally seen in high current flyback design using passive rectification.

CIRCUIT DESCRIPTION

The PCB design is basically optimized as a test platform to evaluate of active rectification using Smart synchronous rectification and as well as basic features of flyback converter operating in quasi-resonant mode.

This board has 2-pin connector ( CON1 ) for AC input and a time-lag type 3.5A fuse for input current overload protection. Minimum input filtering is provided (Cp1-Xcap) before AC input voltage (90-264VAC) is routed to a 6Amp-bridge rectifier (DB1).

Primary side controller (U2) basically drives the primary Mosfet Q1 to operate in Critical-Conduction mode to eliminate turn-ON switching loss thru ZVS (zero voltage switching only occurs when NVsec > Vdcin ) or thru LVS ( low-voltage switching when nVsec< Vdcin) to reduce capacitive losses of Q1 especially at high line condition. The switching frequency Fsw at full load varies from ~38 to ~76kHz typically from low to high input condition and falls back to minimum value (fixed ~ 6 -10kHz) to reduce input power during light load condition.

Auxiliary winding is loosely monitored by demagnetization pin4 of U2 through Dp3, Rp5 and Rp11 network that sets the OVP limit with Rp6 and Rp11 sets the over power limit of the converter.

Resonant capacitor Cp7 is added to augment the overall parasitic winding capacitance and the primary mosfet Q1’s Coss to achieve ZVS and LVS at low and high input line condition respectively.

Optocoupler U3 provides isolated output voltage feedback to the primary side. The output voltage level across load connector CON2 (+16Vo) is monitored and regulated by the V/I Secondary error amplifier U4 (AQ105 or AS4305) that also manages the output current limiting function by monitoring the voltage across the RS25-26 current sense resistors.

The power stage of the secondary is using 2-SO8 low IRF7853 synch-fets (SR) in parallel to implement the low-side synchronous rectification. In this configuration, it is simpler to derive the Vcc supply for the U1 (IR1166 SO8-IC) controller directly from the DC output Vout. Jumper J5 is used to isolate U1’s Vcc from Vout so that user may easily evaluate IC’s power consumption especially during standby load condition. In the absence of a sensitive low current probe, the quiescent current Icc through Dp4 can be calculated from the differential voltage across the Rs17. The decoupling capacitor Cs17 and Cs18 provides additional filtering which is necessary to clean high frequency noise especially when U1 is driving several mosfets (SR1 // SR2) with high Qg parameters normally associated with high currentlow voltage mosfets.

The Vd and Vs sense pins monitor the voltage (Vsd) across the sync rect mosfets and proper attention was taken during PCB routing to ensure the integrity of differential voltage Vsd. This is done by directly taking the signal Vd from the drain pins of SR1//SR2 using a dedicated trace.

Probe points as well as redundant test hook points are provided to facilitate easy probing of essential test waveforms.

Switch Mode Power Supply 12V 8A

The UCC3807 family of high speed, low power integrated circuits contains all of the control and drive circuitry required for off-line and dc-to-dc fixed frequency current mode switching power supplies with minimal external parts count.

Click on the picture to enlarged

These devices are similar to the UCC3800 family, but with the added feature of a user programmable maximum duty cycle. Oscillator frequency and maximum duty cycle are programmed with two resistors and a capacitor. The UCC3807 family also features internal full cycle soft start and internal leading edge blanking of the current sense input.

The UCC3807 family offers a variety of package options, temperature range options, and choice of critical voltage levels. The family has UVLO thresholds and hysteresis levels for off-line and battery powered systems. Thresholds are shown in the table below.

The circuit shown above illustrates the use of the UCC3807 in a typical 100-W, 200-kHz, universal input forward converter produces a regulated 12VDC at 8 Amps. The programmable maximum duty cycle of the UCC3807 allows operation down to 80VRMS and up to 265VRMS with a simple RCD clamp to limit the MOSFET voltage and provide core reset. In this application the maximum duty cycle is set to about 65%. Another feature of the design is the use of a flyback winding on the output filter choke for both bootstrapping and voltage regulation. This method of loop closure eliminates the optocoupler and secondary side regulator, common to most off-line designs, while providing good line and load regulation.

DC Servo Amplifier with Negative Feedback

And negative feedback amplifier voltage than current negative feedback amplifier as excellent transient nonlinear distortion and intermodulation distortion characteristics, the frequency amplifier curve flat, high and low frequency response more exhibition wide; more important is that circuit Will load impedance into the feedback network, it can change the speakers of such fierce resistance to the load compensation, coupled with stable and reliable performance, than the negative feedback voltage amplifier has more advantages, the current negative feedback current amplifier is widely For the modern high-fidelity audio amplifier.



Circuit above is an excellent performance, improve the design of the fever-100 W × 2 DC Servo Amplifier current negative feedback stereo amplifier, formed by the two TDA7294, the frequency response of 10 Hz ~ 100kHz. The use of sophisticated audio Yun-double as the two-channel amplifier DC Servo Amplifier output. Speakers from the protection of ASIC μPC1237HA driver completed the relay switch-mute and amplifier output DC offset protection, and other speakers. When the AC power plug, the relay will be delayed for some time speakers access amplifier; disconnecting the AC power when, μPC1237HA detected exchange loss, immediately disconnect the speaker to relay, the amplifier is the complete elimination of the set, Shutting down the transition process the impact of noise on the speakers.

In actual use, taking into account the electricity grid fluctuations Rectifier amplifier output voltage ± Vs volatile, in order to avoid over-voltage and high temperature in the state of damage TDA7294 (Note pressure in the temperature of 25 ℃ under the conditions, if the temperature exceeds 25 ℃, TDA7294 the value will subsequently reduce the pressure), the exchange recommended power supply voltage transformer CT-AC26V × 2.

Power Supply Circuit 12-15 Volt 20A

Output voltage of the power supply circuit is adjustable from fine potensiometer from 12V to 15v. It is suitable for all 12V power supply devices, or devices which are normally connected to a 12V battery or a vehicle with a 12V power supply system. This tension is usually 13.8 V.



For above reason, The Power Supply is also set to this tension, all right, however, any voltage from 12V to 14V. In this case, the tension is set somewhere around 13.6 V. To provide tension resistance in addition to voltage regulator 78S12. Instead potentiometer 100R inserted resistor 56R.



The scheme of the power supply is simple, but it is partly taken from some of the schemes taken up in the past. The material used is easily obtainable in electronic component shops, and this was the condition when I started to design this power supply.

Bench Power Supply Unit

Here is a rather novel Power Supply Unit (PSU) for the workbench which can deliver 0-15 volts and up to 0-3 amperes. I have not shown any mains fuses as this I will leave to your own devices. You should as a minimum have a fuse in the primary and secondary of T1.


The circuit is a little cramped, but this is because I will also be posting the circuit to QRP@WW on the packet radio system. There are two resistors in the circuit marked "*see text" and these set the maximum output current limiter to a suitable value. Omit the 0R2 resistor and the PSU will deliver 100mA maximum. In this event the output TR8 can be almost any medium power transistor such as the 2N3053, BFY51 etc. It must be mounted on a heatsink no matter what output you need.

The 6R8 resistor may be an ordinary 1/4 watt resistor and is used on all versions of the PSU. The additional 0R2 (0.2 ohms) sets the maximum current limiting to about 3 amperes. If you only want a 0-1 ampere PSU then use 0.68 ohms. T1 is a simple transformer suitably rated for the output power you want and having a 15 volt secondary. This should give over 21v of DC when rectified (measured at D1 cathode - ground). The anode of D7 should also have -21 volts with respect to ground.

If you have problems getting a 15 volt transformer then you can use a 12-volt transformer and add a few extra turns to the secondary. Most transformers have enough room for you to thread another 20 turns of wire through the former to put in series with the secondary winding. By sure to use the same thickness of wire as used for the factory made secondary.

D1, D2, D3, D4, D5, D6, D7 and D8 are all 1N4001 or better for up to 500mA but 1N5401 should be used if you want up to 3 amperes. D9 and D10 may be almost any small silicon diode such as 1N4148 etc. The 1K0 potentiometer in parallel with D9 is the current set pot. The 47K pot is the output voltage pot. With the values shown you should be able to get a range of 0v to a little over 15v.

HOW IT WORKS

TR4 and TR5 compare the ground (0v) with the wiper of the 47K pot which should also be 0v. If the voltages are not equal then TR5 controls the output darlington (TR6 and TR7) until the voltages become equal. If the 47K pot is set to maximum (top) then the output voltage of the PSU must be zero in order to get a balance. If the output voltage pot is set to the bottom then the voltage balance will only occur if there is +15 volts on the top of the 47K pot; the other end of the 15K being -5v.

In the event TR7 draws too much current then the voltage drop accross the 6R8 resistor will turn on TR1 which in turn turns on TR2. TR2 sinks the 0v reference to -20 volts. The TIP41 is incapable of delivering a negative output voltage so the output remains at 0v. The base of TR1 is "lifted" by up to 0.7v due to the current flowing through D10. This overcomes the TR1 base-emitter voltage drop needed to turn it on. You can adjust the 22K resistor a little so that the current control goes all the way down to 0mA and no further. 5K6 is about the absolute minimum resistance you should use.

The power supply itself is very simple and is nothing more than a full-wave bridge rectifier. The AC from the transformer is also capacitively coupled to yet another rectifier to provide the negative voltage required to allow the PSU to be controlled down to 0 volts. TR3 regulates this to stabilise the negative reference voltage.

POSSIBLE PROBLEMS

In the event of self-oscillation of the PSU then under no circumstances put any capacitance accross any signal in the control path. This will just make the problem worse. The cure for this would be to add capacitance between the base/emitter of the controlling transistors TR4 and TR5.

6V to 12V Converter

This inverter circuit can provide up to 800mA of 12V power from a 6V supply. For example, you could run 12V car accessories in a 6V (British?) car. The circuit is simple, about 75% efficient and quite useful. By changing just a few components, you can also modify it for different voltages.


Parts

• R1, R4 ------ 2 2.2K 1/4W Resistor
• R2, R3 ------ 2 4.7K 1/4W Resistor
• R5 ------ 1 1K 1/4W Resistor
• R6 ------ 1 1.5K 1/4W Resistor
• R7 ------ 1 33K 1/4W Resistor
• R8 ------ 1 10K 1/4W Resistor
• C1,C2 ------ 2 0.1uF Ceramic Disc Capacitor
• C3 ------ 1 470uF 25V Electrolytic Capcitor
• D1 ------ 1 1N914 Diode
• D2 ------ 1 1N4004 Diode
• D3 ------ 1 12V 400mW Zener Diode
• Q1, Q2, Q4 --- 3 BC547 NPN Transistor
• Q3 ------ 1 BD679 NPN Transistor
• L1 ------ 1 See Notes
• MISC ------ 1 Heatsink For Q3, Binding Posts (For Input/Output), Wire, Board

Notes

• L1 is a custom inductor wound with about 80 turns of 0.5mm magnet wire around a toroidal core with a 40mm outside diameter.
• Different values of D3 can be used to get different output voltages from about 0.6V to around 30V. Note that at higher voltages the circuit might not perform as well and may not produce as much current. You may also need to use a larger C3 for higher voltages and/or higher currents.
• You can use a larger value for C3 to provide better filtering.
• The circuit will require about 2A from the 6V supply to provide the full 800mA at 12V.

12VDC Fluorescent Lamp Driver

A number of people have been unable to find the transformer needed for the Black Light project, so I looked around to see if I could find a fluorescent lamp driver that does not require any special components. Here it is. It uses a normal 120 to 6V stepdown transformer in reverse to step 12V to about 350V to drive a lamp without the need to warm the filaments.


Parts

• C1 ----- 1 100uf 25V Electrolytic Capacitor

• C2,C3 -- 2 0.01uf 25V Ceramic Disc Capacitor
• C4 ----- 1 0.01uf 1KV Ceramic Disc Capacitor
• R1 ----- 1 1K 1/4W Resistor
• R2 ----- 1 2.7K 1/4W Resistor
• Q1 ----- 1 IRF510 MOSFET
• U1 ----- 1 TLC555 Timer IC
• T1 ----- 1 6V 300mA Transformer
• LAMP --- 1 4W Fluorescent Lamp
• MISC --- 1 Board, Wire, Heatsink For Q1

Notes

• Q1 must be installed on a heat sink.
• A 240V to 10V transformer will work better then the one in the parts list. The problem is that they are hard to find.
• This circuit can give a nasty (but not too dangerous) shock. Be careful around the output leads.