Radio Transmitter

For your comments, suggestions, or your looking for a circuit or design, or a electronics designer, please send you Inquiries to pcb1001@gmail.com. We are happy to help and attend to your concerns.

Tuesday, April 28, 2009

High Gain Amplifier

The amp is based on the High Gain PCB, so uses a pair of LM3876 (or LM3886) power opamps, run from a ±35V supply. I used a cut-down P88 preamp PCB because I only wanted one preamplifier stage, but the entire board can also be used. Alternatively, the P19 amp can be run at higher gain than normal, alleviating the need for a preamp at all. The down side of this is that the noise level will be higher, and background noise may be audible with efficient speakers and/ or very quiet surroundings.


The internal layout can be seen best in Figures 2 and 3. The main heatsink runs down the middle of the amp, and it separates the input and output stages. The material is 10mm thick aluminium, 45mm high and 180mm long. Because this is a prototype of the chassis assembly, there are several things that I would do differently if I build another. The chassis is more complex than it should be, and there are several opportunities for simplification. These became obvious after the basic chassis was well underway (naturally), and there were holes that I couldn't 'undrill' to simplify construction. Such is life.


The front top view shows the general layout of the amp's internals. On the left is the sheet aluminium clamp that holds the capacitors in place, and against the central heatsink section is the P19 amp board. On the other side of the heatsink is the input selector switch and then the ½ P88 board.

Along the rear (from left to right) is the DC connector, speaker outputs and inputs. As it turns out, 4 inputs is enough for my application, and had I restricted it to that the shield between the last set of inputs and the speaker connectors would not have been needed.

The DC connector, speaker connectors and input RCA sockets are all mounted on blank fibreglass PCB material to insulate them from the chassis. Where needed, the copper was removed to create a rudimentary PCB pattern - this is evident on the DC and speaker panels. The boards were 'etched' using a rotary tool (Dremmel or similar). Although the resolution and accuracy are not good enough for an amplifier, this method works very well for such applications.


The back view shows the vent slots along the top, and you can see that the RCA connectors do not contact the chassis. Naturally, the speaker terminals are insulated. The DC connector is clearly visible on the right. It is a lot easier to simply make the back panel a little shorter than the other panels than it is to cut slots as shown. Even with a milling machine, these are somewhat tedious to do, and it is difficult to get perfect alignment without proper jigs. The hole for the DC plug and socket is relatively easily made using a drill and square file. The switch hole will require some fairly tedious filing if you use a rectangular switch as shown, however you can use any switch at all, because it only has to switch 9V AC.

Again, the slots look cool, but a series of holes will work just as well. There are a number of other refinements as well, and these are listed in the construction section below.

The Electronics

As noted above, the electronics are based on two existing projects - P19 stereo 50W amplifier, and P88 high quality preamp. The schematic is shown below (one channel only), and the P88 only uses the second half of the PCB. The P19 power amp is constructed normally, and there are no changes from the published project.

The inputs can be designated with whatever you want, and you can add more if desired (within the limits of the rear panel real estate). It is important that the gain of the preamp section is kept low enough to ensure that none of your inputs will clip the opamp. Assuming that CD/ DVD players are capable of about 2V, this means that the gain must be kept below 6.5 (16dB). This is not a problem unless you change the values of R7A, B and C, since the maximum gain is limited to about 9.5dB with the values shown.

The caps before and after the volume control can be bypassed completely (using wire links), but I do not recommend that you do so. If there is DC across the pot,it will become noisy and scratchy after a while. Even small amounts of DC can cause problems.

Power Supply Module

The power supply I used is probably overkill, but I simply used parts I had on hand. The schematic is shown below. Although I used zeners for the opamp supply as shown, some constructors are bound to be uncomfortable with such a simple arrangement. The P05 board can be used to provide full regulation, but with only one dual opamp, I'm not sure it is warranted.
A photo of the cpmplete module is shown below. The soft start isn't really needed with a 160VA transformer, but it does no harm, and allows remote low voltage switching. Since this was a requirement (the connectors are illegal for use with hazardous voltages), it was a small price to pay. Although the transformer is happy without the soft start, there is a total of 20,000uF on each supply rail, and this would place great stress on the bridge rectifier.


The two 2.2k 1W resistors across the filter caps in the supply box ensure that the caps will discharge even if the amplifier is not connected. They are not strictly needed, but are recommended to prevent nasty sparks is the amp is connected while the caps are still charged. Large electros can easily maintain a respectable charge for many hours.

The power supply is conventional in almost all respects. I used a 160VA transformer, a 400V 35A bridge rectifier, and a total of 20,000uF per supply rail - 4 x 10,000uF caps in all. When the connecting cable resistance is added in, there is almost no ripple at all at the amplifier, even with both channels at full power. The cable resistance aids filtering, but at the expense of slightly reduced maximum continuous power. I obtained over 40W per channel with both channels driven into an 8 ohm load, and peak short term power is over 60W / channel.

You can use less capacitance of course, but with some increase in ripple and (perhaps) noise. For an amp of this nature, I expect that few constructors will want to use less than about 4 x 4,700uFcaps. Additional capacitance can also be used in parallel with the zener diodes, but 100uF 16V caps fit the P88 board easiliy. There is nothing to suggest that more capacitance will serve any purpose.

Since the amplifier is absolutely dead quiet even at full volume with unterminated inputs, there is nothing one can do to make it any better. Placing one's ear right next to the speaker (one of average sensitivity), circuit noise is just audible. There is no hum at all.

Sunday, April 26, 2009

Super Amplifiers 300W Output Power

The circuit described on this page is a modification of the original Double Barreled Amplifier. The circuit has been simplified somewhat. The circuit board layout is smaller and much more compact. The driver transistors now mount on the circuit board instead of on external heat sinks. And the circuit has the feedforward compensation that I describe for the Low TIM Amplifier.

The original circuit board for one channel had eight 5-watt resistors on it, one in series with the emitter of each output transistor. On the new layout, four of these have been moved to the heat sink channel where they solder between pins of the transistor sockets. This change not only helps make the circuit board smaller, but it eliminates eight wires between the heat sink and the circuit board. One of the figures below illustrates how these resistors are installed in the heat sink channel.



If you build this amplifier, you must keep the wiring between the heat sinks and the circuit boards as short as possible if you don't want oscillation problems.

When you test the circuit boards before connecting the power transistors, temporarily connect a 10 ohm resistor in series with a 0.1 ufd capacitor from the loudspeaker output to the power supply ground.

The Circuit Boards





We do not have circuit boards for the Double Barrelled Amplifier. If you wish to build it, you must make your own. Two drawings show the parts layout on the board, one with circuit traces and one without. These are scaled by a factor of 1.5. The other shows the circuit traces only. All layout views are from the component side of the board. You must flip the layout for the foil traces over to obtain the solder side view. The circuit board measures 4 inches by 6 inches. To my knowledge, there are no errors in the layout. If you decide to use it, you should carefully check it for errors because I could have easily made a mistake.


We do not recommend that you make the circuit boards unless you have experience in doing it. A source of materials for making your own printed circuits can be found here. I have been told that their "Press and Peel Blue" product (not the wet stuff they sell) can be used to successfully make boards with traces as narrow as 0.01 inch. The smallest traces on the amplifier layout are 0.03 inch wide. The PnP Blue product is basically a transfer medium that allows you to transfer the toner image from a laser printer directly onto bare copper clad board and then etch it in FeCl3 (ferric chloride).


After you etch the board, the copper should be cleaned with steel wool, lightly coated with solder flux, and then "tinned" with a soldering iron and rosin core solder. Do not use a commercial tinning solution that you dip the board into. It is almost impossible to solder a board that is tinned with one of these products because they corrode very quickly. When you drill the board, you should use the correct size drill bit for the pads. The hole diameters I recommend are: small pads - 0.032 inch, medium pads - 0.040 inch, large pads - 0.059 inch, mounting holes - 0.125 inch. If you do not use a sharp drill bit, you can pull the pads off the board when you drill it.


Circuit Description

If you compare the Double Barreled circuit to the Low TIM circuit, you will see a lot of similarity between the two. Indeed, there is a Low TIM Amplifier embedded in the Double Barreled Amplifier. The major difference between the two is that transistors are added in series with those in the Low TIM circuit to form the Double Barreled circuit. By doing this, the voltage across the transistors is decreased so that the power supply voltage can be increased for higher output power.

Basically, the circuit description for the Low TIM Amplifier also applies to the Double Barreled Amplifier. The major difference between the two is the addition of transistors Q22 through Q31. Q22 is connected as a common base stage at the output of Q12. The two transistors form a cascode stage. The base of Q22 connects to the junction of R52 and R54. These two resistors are equal and are connected as a voltage divider between the loudspeaker output and the positive rail. This forces the base voltage of Q22 to float half way between the loudspeaker output voltage and the positive power supply rail. Similarly, Q13 and Q23 form a cascode stage. R53 and R55 force the base of Q23 to float half way between the loudspeaker output voltage and the negative power supply rail. The addition of Q22 and Q23 cause the collector to emitter voltages of Q12 and Q13 to be approximately one-half of what the voltages would be without Q22 and Q23.

Transistors Q24 and Q25 connect in series with the pre-driver transistors Q14 and Q15. The base of Q24 floats half way between the output voltage and the positive rail. The base of Q25 floats half way between the output voltage and the negative rail. The addition of Q24 and Q25 cause the voltages across Q14 and Q15 to be approximately one-half of what they would be without Q24 and Q25. Similarly, transistors Q26 through Q31 cause the voltages across Q16 through Q21 to be approximately one-half of what they would be without Q26 through Q31. By connecting the transistors in series in this way, the rail voltages can be increased for higher output power.

The basic construction details of the Low TIM Amplifier also apply to the Double Barreled Amplifier. There are two short circuit jumper wires that must be soldered on the circuit board. These are marked with a J on the layout. In addition, you must solder a short circuit jumper in place of C6B if you use a non-polar capacitor for C6A. This is explained in the parts list for the Low TIM Amplifier. Because there are eight output transistors, two main heat sinks per channel are required. Q18, Q20, Q28, and Q30 should be mounted on one and Q19, Q21, Q29, and Q31 on the other. Resistors R61 through R64 and wires connecting the collectors of Q18 and Q20 and the collectors of Q19 and Q21 mount on the heat sinks. These connect between lugs on the transistor sockets. The four bias diodes D1 through D4 can be mounted on either heat sink. It is not necessary to divide the diodes between the two heat sinks because both heat sinks will operate at the same temperature. I recommend setting the voltage across Q7, i.e. the voltage between the collectors of Q22 and Q23, so that that amplifier is biased at 120 mA. This will give the same quiescent power dissipation per heat sink as in the Low TIM Amplifier.

Testing the Circuit Boards

After you solder the parts to the circuit board, it is tested using the same procedure specified for the Low TIM circuit board. First, you must solder the short circuit jumper across Q7 and you must solder the 100 ohm 1/4 W resistors from the loudspeaker output to the emitters of Q16 and Q17. If you don't have a bench power supply that puts out plus and minus 85 to 93 V dc, you can test the circuit board at a lower voltage. I would prefer test voltages of at least plus and minus 50 V dc. An option is to connect bench power supplies in series to obtain the plus and minus 85 to 93 V dc. I have routinely connected two 40 V Hewlett Packard power supplies in series with the positive and negative outputs of a Hewlett Packard 50 V dual power supply, and I have never had any problems. To protect the circuit boards, you might want to put a 100 ohm 1/4 W resistor in series with the plus and minus power supply leads for the tests. The current drawn by the circuit should be low enough so that the voltage drop across these resistors is less than 1 V if nothing is wrong on the circuit board. There are 2 ground wires from the circuit board. Both must be connected when testing the boards.

I can't stress how important it is to be careful in testing a circuit board. Even simple errors can cause the loss of many expensive transistors. I always use current limited bench power supplies to test a circuit board before and after connecting the power transistors. I also bias an amplifier using current limited power supplies in place of the amplifier power supply. When I initially power up an amplifier with its own power supply, I always use a Variac variable transformer to slowly increase the ac input voltage from 0 to 120 V rms while observing the amplifier output on an oscilloscope with a sine wave input signal. If I see anything wrong on the oscilloscope, I turn the Variac to zero and try to diagnose the problem using the bench power supply. I never use a load on the amplifier for these tests.

Parts List

With the following exceptions, the parts for the Double Barreled Amplifier are the same as for the Low TIM Amplifier.

Capacitors

  • C10, C11 - 15 pF mica
  • C13, C14 - 100 uFd 100 V radial electrolytic
  • C21, C22 - 47 uFd 100 V radial electrolytic
  • C26, C27 - 270 pF mica
  • C28 - 0.01 uFd 250 V film

Transistors

  • Q1, Q2, Q5, Q7, Q9, Q10 - MPS8099 or MPSA06
  • Q3, Q4, Q6, Q8, Q11 - MPS8599 or MPSA56
  • Q23, Q24 - 2N3439
  • Q22, Q25 - 2N5415
  • Q26 - MJE15030
  • Q27 - MJE15031
  • Q28, Q30 - MJ15003
  • Q29, Q31 - MJ15004

Diodes

  • D5, D6 - 1N4934 fast recovery rectifier
  • D13 through D16 - 1N5250B 20 volt zener diode

Resistors

  • R13, R14 - 5.6 kohm 1 watt (This value is for 85 V power supplies. For other power supply voltages, the formula is on the Parts List page for the Leach Amp.)
  • R28, R29 - 200 ohm 1/4 watt
  • R30, R31 - 3.9 kohm 1 watt
  • R37 through R40 - 470 ohm 1/4 watt
  • R41 through R44 - 10 ohm 1/2 watt (changed 6/27/00)
  • R52 through R55 - 6.2 kohm 1 watt
  • R56 through R59 - 10 ohm 1/2 watt (changed 6/27/00)
  • R60 - 39 ohm 1/4 watt
  • R61 through R64 - 0.33 ohm 5 watt. These 4 resistors are mounted on the heat sinks between solder lugs on the power transistor sockets. The wires that connect the collectors of Q18 and Q20 and the collectors of Q19 and Q21 are also soldered between the lugs on the sockets. Keep all leads as short as possible and use insulation stripped from hookup wire around the bare leads of the resistors.
  • R65, R66 - 300 ohm 1/4 watt

Heat Sinks

  • Double the number of heat sinks required for the Low TIM Amplifier.

Power Supply Components

The power supply circuit diagram is the same as for the Low TIM Amplifier. The parts are the same with the following exceptions.

  • T1 - The transformer should have either a center tapped secondary or two separate secondary windings which can be wired in series. With 120 V ac rms applied to the primary, the no load secondary voltage should be 120 to 130 V ac rms for a center tapped secondary or 60+60 (60x2) to 65+65 (65x2) V ac rms for two secondary windings. This should give a no load amplifier power supply voltage of plus and minus 85 to 93 V dc. Some transformers are rated at 115 V ac rms on the primary. With 120 V ac rms applied, the secondary voltage will be greater by a factor 120/115. If the transformer is rated at full load, its no load voltage will be 15% to 20% higher. I would recommend a transformer current rating of at least 6 A. The transformer I used in each of my two original Double Barreled Amplifiers was the Signal 230-6. It had two center tapped 115 V 6 A secondaries which I wired in parallel to obtain a secondary rating of 115 V at 12 A. The primary had three voltage taps: 105 V, 115 V, and 125 V. I wired the AC line input to the 115 V tap. With 120 V AC applied to the 115 V tap, I got plus and minus 85 V DC on the power supplies and 270 W into an 8 ohm load. If I had used the 105 V primary taps, the power supply voltage would have increased to about 93 V and the amplifiers would have put out over 300 W. The Signal transformer was definately an overkill. It weighed 38 pounds. But it would really kick you know what. To my knowledge, this transformer now is available only by special order.
  • C1P, C2P - I used two Mallory CG832U100G1 8,600 uFd 100 V capacitors in parallel for each of these so that I had 34,400 uFd total in each of my two amplifiers. This was probably an overkill. The energy stored in the eight apacitors was about 250 joules. This is enough energy to lift a 25 pound dog over 7 feet off the floor. For C1P and C2P, I would recommend at least 10,000 uFd total for each. The voltage rating should be 100 V or greater.

New Vacuum Tube Amplifier


Features

  • Output: 4-6550's in triode-mode class AB2 push-pull parallel. About 80 watts RMS per channel.
  • No global negative feedback. Several local loops with limited negative feedback.
  • Ultra-wide bandwidth Plitron toroidal output transformer.
  • Servo to maintain precise dc-balance in the output circuit.
  • MOSFET-regulated power supplies. Relative rather than absolute voltage reference.
  • Designed using extensive PSpice computer simulation.
  • Constructed as a pair of monoblocks.

Schematic diagram


Notes on the schematic diagram

  • The power supply has been simplified-- Power transformers and rectifiers have been omitted and some parts have been omitted from the MOSFET voltage regulator circuits: 1N5242 zener diodes between the source and gate and 10k resistors in series with the gate. These parts serve as protection in case of accidental short circuits, but don't affect the operating point. The full power supply schematic is shown below.
  • 6SN7's are used instead of 12AU7's for the driver tubes. They have the same plate characteristics, but they have higher maximum plate voltage (450 vs. 330 V) and greater plate dissipation (3.75 vs. 2.75 W per section). Think of the octal-based 6SN7 as a 12AU7 on steroids.
  • The NODESET blocks (lower right) initialize dc levels in the bias servo so the simulation runs properly. They don't exist as physical entities.
  • All resistors are 1/2 watt unless noted, and with the following exceptions. R3P, R4P and R5C through R8C (68K) are 2W. R9C through R12C (20 ohm) are 1W: I used 2-10 ohm resistors in series to make them. R9S through R12S (200 ohms) are 2W.
  • Capacitor voltage ratings: It never hurts to go over the minimum, though the capacitors will be larger and may cost more. I often use caps with higher voltage ratings because I have them on hand or found them at a good price in an electronics surplus shop. Here are some minimum ratings: C1G: 100V. C2G: 400V (600V would be better; necessary without the time delay); C3G and C4G: 400V (600V would be better); C3M and C4M: 400V; C5G through C8G: 600V; CBS2, CBS4, CBS6 and CBX1 through CBX3: 100V. The voltage ratings of many of the power supply capacitors are shown on the circuit board wiring diagrams, below.
  • Some of the feedback connections may be a little hard to trace. ORN goes between the 20 ohm output transformer secondary feedback winding and R3F near TU3. Similarly, VLT goes to R4F near TU4. BLU and BLK on the output transformer secondary speaker winding go to the output tube cathode circuits.
  • RLS (5 ohms) is a simulated speaker load.
  • CRF: All three nodes are connected together.
  • BRN and VIO are not used. They are the ultra-linear taps. In the original version of TENA there was a switch to select output tube screen grid connections between BRN and VIO (UL mode) and GRN and YEL (triode mode).

Input stage/phase inverter

Input stage TU1 is a simple voltage gain stage with local negative feedback, derived from the R1B, R1C voltage divider. It is capacitively coupled to split load phase inverter TU2. The capacitor has an unusually low value-- 0.01 µF-- because TU2 has an exceptionally high input impedance-- several Megohms. The advantage of capacitive coupling is that it allows the voltage level in TU2 to be set for maximum output and it allows the ac current in TU2 to be precisely equal to, but 180 degrees out of phase with, the current in TU1. The net ac current drawn by these two tubes from V+420 is therefore zero. This is an effective way of isolating the audio signal from the power supply, which doesn't need to supply ac current. In conventional designs ac signal often has to flow through electrolytic capacitors, which are grungy leaky devices with memory-- harmful to audio quality. I designed TENA to draw zero net ac current from all power supply outputs (easy to do in a push-pull design), at least up to the power level where one of the output tube pairs starts cutting off.

Toroidal output transformer

We chose the Plitron toroidal transformer because of its exceptional bandwidth: -3 dB at over 200 kHz, the result of high primary inductance (the good stuff) and low leakage inductance (the bad stuff-- kind of like HDL and LDL cholesterol)-- much better than can be achieved with a conventional EI transformer. High bandwidth is important because output transformers have an intrinsic second order rolloff, which can make them unstable in the presence of negative feedback unless careful phase compensation is applied (see Feedback and Fidelity). Phase compensation reduces the bandwidth, which is not a problem with the Plitron toroidal transformers. But this bandwidth comes at a price-- toroidal transformers are much less tolerant of dc-imbalance than EI transformers; they may saturate at dc imbalances as low as 8 mA. (I don't know the exact number; I never simulated it.) You would have to set the bias of each tube individually, and then you'd have to worry about how the tubes age. So I designed a bias servo circuit to maintain perfect dc-balance under all circumstances except outright tube failure.

The Plitron PAT 4006CFB 100 Watt toroidal output transformer is not currently listed on Plitron's website, but I've heard (June 2003) that it is available. Contact Norman Woo. The closest models are the 4006, which lacks the special feedback winding, and the 2100-CFB which has a higher primary impedance. The minimum feedback version of TENA (below) works with the 4006.

Bias servo and adjustment

The time-averaged (low pass filtered) dc current of an output tube operating in class AB fixed bias is relatively constant at low power levels but increases at high power levels. For this reason a fixed voltage cannot be used as a reference for biasing the output tubes. One tube (TU9, driven by TU5) operates at fixed bias, and its low pass filtered cathode voltage (CRF) is used as the reference for biasing the other tubes.

The bias servo is illustrated in the lower left of the schematic. It uses the LM324 quad op amp-- cheap but perfectly adequate. Inputs U1A, U1B and U1C of the LM324 compare cathode voltages 10C, 11C, and 12C with reference voltage CRF, which is the voltage on cathode 9C low pass filtered with RBS2 = 33k and CBS1 = 10µF ( located near U1B on the schematic). The LM324 outputs control the P-channel MOSFETs, each of which controls a voltage divider between VBB (-90V) and VOP (+12.5V) to deliver the appropriate bias voltage to the driver grid circuits (BIAS_6, BIAS_7, and BIAS_8). This measures between -45 and -50V in my amplifiers, which operate at 60 mA plate current. Audio purists please note: the servo operates at extremely low frequencies; the op amp and MOSFETs are well outside the audio signal path.

A single potentiometer, RB5 (in the VBB supply, bottom center), controls the bias current directly in TU9, and all the other tubes indirectly through the servo. Bias current may be measured across any of the 20 ohm resistors R9C-R12C as E/20. They should all be the same if the servo is working properly. 1 to 1.2 volts is a good nominal value, corresponding to 50 to 60 mA per tube (70 mA was used in the Dynaco Mark III). Increasing the current increases power consumption and reduces tube life and output power, but moves you closer to Class A (where both tubes always conduct).

Class AB2 output stage and drivers

Class AB2 differs from the more common class AB1 in that the output stage grid is driven positive-- it draws grid current-- at high power levels. Class AB2 has no advantage for output tubes operating in pentode mode and little advantage for ultra-linear mode. But it results in a huge power boost for output tubes operating in triode mode. You can get almost as much power out of class AB2 triodes as you can out of class AB1 pentodes.

If you try to do operate in class AB2 with conventional capacitive coupling, the coupling capacitor starts charging as soon as grid current is drawn. This drives the grid negative-- toward cutoff, and it recovers with the RC time constant of the coupling capacitor and grid resistor. To operate successfully in class AB2, the output stage must be either transformer or direct coupled. I chose direct coupling because interstage transformers are expensive and have limited bandwidth.

The direct coupled drivers are the source of much of TENA's complexity. Because the quiescent grid voltage of each output tube must be set individually to control its quiescent (dc) current, one driver tube (TU5-TU8) is required for each output tube (TU9-TU12). Cathode followers (CF's) were chosen because they have low output impedance and can source the needed output tube grid current. The cathodes have to be somewhere near -50V to properly bias the output tubes. This means the CF must be driven by voltages outside the range of conventional power supplies, hence the need for VDR- and VDR+: the price of perfection. In reviewing the design I find that the driver tubes may be operating a little too conservatively-- dissipating only 0.78 W (of a 6SN7 maximum of 3.75 W). I've discussed driver dissipation under PSpice output, below. I may increase VDR+ from 205 to around 250 V by increasing RD1 from 470k to 680k. This would reduce the power dissipation in MOSFET MD1.

Output tube grid stop resistors R9G-R12G play an important role in TENA's soft clipping. When power levels become high enough level for grid current to be drawn, a voltage drop across these resistors gradually limits the plate current. Soft clipping consists of low order harmonics which have much less adverse effect on sound quality that the high order harmonics characteristic of abrupt clipping. But total harmonic distortion for soft clipping amplifiers tends to be higher. Yes, lower harmonic distortion doesn't mean better sound. See "The great harmonic distortion scam" in Feedback and Fidelity. TENA oscillated when the grid stop resistors were removed. This was the only performance feature PSpice didn't catch. The reason is that the output transformer model is somewhat simplified-- it's extremely difficult to model its distributed capacitance.

Power supplies

The time delay circuit (U3 (the 555B chip), Q1, Relay_SPDT_nb, RT1, CT1, CT2, RT3, D1, RV1, and RT4) has apparently never been implemented. RT4 should be replaced by a straight wire; VBIN is connected directly to NTC (negative temperature coefficient; 50 ohms cold; Mouser527-3504-50) thermistor RV10.

The precise values of most of the capacitors in the power supply, particularly CV1, CV2, CB1, CB2, CD1 and CD2, are not critical. In many cases they were determined by parts availability. If the values are 2 uF or under they are film capacitors. If they are over 2 uF they are electrolytics.

Depending how you count there are two (power transformers), four (rectifier circuits) or six (voltage levels). All use fast recovery rectifier diodes. All except VDR- are taken from the mighty Plitron 854710 toroidal power transformer, which I can't seem to find in their catalog. Toroidal power transformers perform well, but they have less of an advantage than toroidal output transformers-- you don't need wide bandwidth for 60 Hz. The CL80 inrush current limiter limits turn-on current in the tube filaments.

Saturday, April 25, 2009

Universal Input Linear Fluorescent Ballast

Features
  • Drives one 35 W TL5 Lamp
  • Input Voltage: 80 VAC to 260 VAC
  • High Power Factor/Low THD
  • High Frequency Operation
  • Lamp Filament Preheating
  • Lamp Fault Protection with Auto-Restart
  • Low AC Line Protection
  • End of Lamp Life Shutdown
  • IRS2166D(S)PbF HVIC Ballast Controller
The Board is a high efficiency, high power factor, fixed output electronic ballast designed for driving rapid start fluorescent lamp types. The design contains an EMI filter, active power factor correction and a ballast control circuit using the IRS2166D(S)PbF Ballast Control IC1.

The Board consists of an EMI filter, an active power factor correction section, a ballast control section and a resonant lamp output stage. The active power factor correction section is a boost converter operating in critical conduction mode, free-running frequency mode. The ballast control section provides frequency modulation control of a traditional RCL lamp resonant output circuit and is easily adaptable to a wide variety of lamp types. The ballast control section also provides the necessary circuitry to perform lamp fault detection, shutdown and auto-restart.

This board is designed for single TL5/35W Lamp, voltage mode heating (JV1 and JV2 mounted, JC1 and JC2 not mounted). TL5 lamps are becoming more popular due to their lower profile and higher lumen/ watt output. These lamps, however, can be more difficult to control due to their higher ignition and running voltages. A typical ballast output stage using current-mode filament heating (filament placed inside L-C tank) will result in excessive filament current during running. The output stage has therefore been configured for voltage-mode filament heating using secondary windings off of the resonant inductor LRES. The lamp has been placed outside the under-damped resonant circuit loop, which consist of LRES and CRES. The filament heating during preheat can be adjusted with the capacitors CH1 and CH2. The result is a more flexible ballast output stage necessary for fulfilling the lamp requirements. The DC blocking capacitor, CDC, is also placed outside the under-damped resonant circuit loop such that it does not influence the natural resonance frequency of LRES and CRES. The snubber capacitor, CSNUB, serves as charge pump for supplying the IRS2166D.

The IRS2166D Ballast Control IC is used to program the ballast operating points and protect the ballast against conditions such as lamp strike failures, low DC bus, thermal overload or lamp failure during normal operations. It is also used to regulate the DC bus and for power factor control allowing high power factor and low harmonic distortion.

Switch Mode Power Supply 100W - 16 at 7A

Generally, Schottky diodes are traditional devices use in passive rectification in order to have low conduction loss in secondary side for switching power supplies. The proliferations of synchronous rectification (SR) idea - which is mostly use in buck-derive topologies - have reached the domain of flyback application in recent years. The use of low-voltage-low-Rdson mosfet has become so attractive to replace the Schottky rectifiers in high current applications because it offers several system advantages such as dramatic decrease in conduction loss and better thermal management of the whole system by reducing the cost investment in heat sink and PCB space.



A number of techniques in the implementation of SR in flyback converters are continuously growing from a simple self-driven (secondary winding voltage detection) to a more complex solution using “current transformer sensing” or combinations of both to improve the existing technology. The idea has become quite complicated though and additional discrete devices have made the cost and part counts issue even worse. Moreover, the issue of reverse current conduction (-due to the delay in sensing the sharp drop of secondary current during turn-off phase of the SR) still lingers on in different input line/ output load conditions. The use of a simple fast-rate-direct-sensing of voltage drop across the mosfet (Vsd) using integrated solution has pave the way for a much simpler and effective means of controlling the SR mosfets as well as alleviating the reverse current and multiple-pulse gate turn-ON issues.

The board is a universal-input flyback converter with single DC output capable of delivering continuous 100W (@ +16V x 6.25A) during active rectification mode. This board is primarily designed to study synchronous rectification using IR1166 in low-side configuration to take advantage of simpler derivation of Vcc supply from converter’s output. It is equipped with necessary jumpers to ease exploring the conduction behavior of synchronous rectifiers SRs in quasi-resonant mode, so discussion would be confined to variable frequency switching in Critical Conduction Mode.

It features the fast Vsd sensing of the IR1166 Smart Rectifier Control IC with gate output drive capability of 1.5Apk. It drives 2 pcs. of SRs in parallel (100V N-ch mosfet IRF7853 in SO-8 package with very low Rdson in its class : 18 mΩ max). This had greatly simplified the overall mechanical design for not having those bulky and heavy heat sinks normally seen in high current flyback design using passive rectification.

CIRCUIT DESCRIPTION

The PCB design is basically optimized as a test platform to evaluate of active rectification using Smart synchronous rectification and as well as basic features of flyback converter operating in quasi-resonant mode.

This board has 2-pin connector ( CON1 ) for AC input and a time-lag type 3.5A fuse for input current overload protection. Minimum input filtering is provided (Cp1-Xcap) before AC input voltage (90-264VAC) is routed to a 6Amp-bridge rectifier (DB1).

Primary side controller (U2) basically drives the primary Mosfet Q1 to operate in Critical-Conduction mode to eliminate turn-ON switching loss thru ZVS (zero voltage switching only occurs when NVsec > Vdcin ) or thru LVS ( low-voltage switching when nVsec< Vdcin) to reduce capacitive losses of Q1 especially at high line condition. The switching frequency Fsw at full load varies from ~38 to ~76kHz typically from low to high input condition and falls back to minimum value (fixed ~ 6 -10kHz) to reduce input power during light load condition.

Auxiliary winding is loosely monitored by demagnetization pin4 of U2 through Dp3, Rp5 and Rp11 network that sets the OVP limit with Rp6 and Rp11 sets the over power limit of the converter.

Resonant capacitor Cp7 is added to augment the overall parasitic winding capacitance and the primary mosfet Q1’s Coss to achieve ZVS and LVS at low and high input line condition respectively.

Optocoupler U3 provides isolated output voltage feedback to the primary side. The output voltage level across load connector CON2 (+16Vo) is monitored and regulated by the V/I Secondary error amplifier U4 (AQ105 or AS4305) that also manages the output current limiting function by monitoring the voltage across the RS25-26 current sense resistors.

The power stage of the secondary is using 2-SO8 low IRF7853 synch-fets (SR) in parallel to implement the low-side synchronous rectification. In this configuration, it is simpler to derive the Vcc supply for the U1 (IR1166 SO8-IC) controller directly from the DC output Vout. Jumper J5 is used to isolate U1’s Vcc from Vout so that user may easily evaluate IC’s power consumption especially during standby load condition. In the absence of a sensitive low current probe, the quiescent current Icc through Dp4 can be calculated from the differential voltage across the Rs17. The decoupling capacitor Cs17 and Cs18 provides additional filtering which is necessary to clean high frequency noise especially when U1 is driving several mosfets (SR1 // SR2) with high Qg parameters normally associated with high currentlow voltage mosfets.

The Vd and Vs sense pins monitor the voltage (Vsd) across the sync rect mosfets and proper attention was taken during PCB routing to ensure the integrity of differential voltage Vsd. This is done by directly taking the signal Vd from the drain pins of SR1//SR2 using a dedicated trace.

Probe points as well as redundant test hook points are provided to facilitate easy probing of essential test waveforms.

Sunday, April 19, 2009

Switch Mode Power Supply 12V 8A

The UCC3807 family of high speed, low power integrated circuits contains all of the control and drive circuitry required for off-line and dc-to-dc fixed frequency current mode switching power supplies with minimal external parts count.

Click on the picture to enlarged

These devices are similar to the UCC3800 family, but with the added feature of a user programmable maximum duty cycle. Oscillator frequency and maximum duty cycle are programmed with two resistors and a capacitor. The UCC3807 family also features internal full cycle soft start and internal leading edge blanking of the current sense input.

The UCC3807 family offers a variety of package options, temperature range options, and choice of critical voltage levels. The family has UVLO thresholds and hysteresis levels for off-line and battery powered systems. Thresholds are shown in the table below.

The circuit shown above illustrates the use of the UCC3807 in a typical 100-W, 200-kHz, universal input forward converter produces a regulated 12VDC at 8 Amps. The programmable maximum duty cycle of the UCC3807 allows operation down to 80VRMS and up to 265VRMS with a simple RCD clamp to limit the MOSFET voltage and provide core reset. In this application the maximum duty cycle is set to about 65%. Another feature of the design is the use of a flyback winding on the output filter choke for both bootstrapping and voltage regulation. This method of loop closure eliminates the optocoupler and secondary side regulator, common to most off-line designs, while providing good line and load regulation.

Saturday, April 18, 2009

DC Servo Amplifier with Negative Feedback

And negative feedback amplifier voltage than current negative feedback amplifier as excellent transient nonlinear distortion and intermodulation distortion characteristics, the frequency amplifier curve flat, high and low frequency response more exhibition wide; more important is that circuit Will load impedance into the feedback network, it can change the speakers of such fierce resistance to the load compensation, coupled with stable and reliable performance, than the negative feedback voltage amplifier has more advantages, the current negative feedback current amplifier is widely For the modern high-fidelity audio amplifier.



Circuit above is an excellent performance, improve the design of the fever-100 W × 2 DC Servo Amplifier current negative feedback stereo amplifier, formed by the two TDA7294, the frequency response of 10 Hz ~ 100kHz. The use of sophisticated audio Yun-double as the two-channel amplifier DC Servo Amplifier output. Speakers from the protection of ASIC μPC1237HA driver completed the relay switch-mute and amplifier output DC offset protection, and other speakers. When the AC power plug, the relay will be delayed for some time speakers access amplifier; disconnecting the AC power when, μPC1237HA detected exchange loss, immediately disconnect the speaker to relay, the amplifier is the complete elimination of the set, Shutting down the transition process the impact of noise on the speakers.

In actual use, taking into account the electricity grid fluctuations Rectifier amplifier output voltage ± Vs volatile, in order to avoid over-voltage and high temperature in the state of damage TDA7294 (Note pressure in the temperature of 25 ℃ under the conditions, if the temperature exceeds 25 ℃, TDA7294 the value will subsequently reduce the pressure), the exchange recommended power supply voltage transformer CT-AC26V × 2.

Motor current control circuit with external power transistors

The L165 is a monolithic integrated circuit in Pentawatt ® package, intended for use as power operational amplifier in a wide range of applications, including servo amplifiers and power supplies. The high gain and high output power capability provide superiore performance wherever an operational amplifier/power booster combination is required.




1.2W AUDIO AMPLIFIER

A very useful audio amp in an 8-pin DIL package. The IC features a very low minimum working supply voltage of 3V, low quiescent current, good ripple rejection, no crossover distortion and low power dissipation. Maximum supply voltages is 16 Volts into 16 Ohms speaker, 12Volts into 8 Ohms and 9Volts into 4 Ohms.



The TBA820M is a monolithic integrated audio amplifier in a 8 lead dual in-line plastic package. It is intended for use as low frequency class B power amplifier with wide range of supply voltage: 3 to 16V, in portable radios, cassette recorders and players etc. Main features are: minimum working supply voltage of 3V, low quiescent current, low number of external components, good ripple rejection, no cross-over distortion, low power dissipation.

Output power: Po = 2Wat 12V/8W, 1.6W at 9V/4W
and 1.2W at 9V/8W.

8 Watts Audio Amplifier

Nice small audio amplifier which use only few parts to give good quality sound. This amp can be used as a simple booster, the heart of a more complicated amplifier or used as a guitar amp. Although not perfect, this amplifier does have a wide frequency response, low harmonic distortion about 1.5%, and is capable of driving an 8 ohm speaker to output levels of around 8 watts with slightly higher distortion. Any power supply in the range 12 to 18 Volts DC may be used.

The TDA 2003 has improved performance with the same pin configuration as the TDA 2002.
The additional features of TDA 2002, very low number of external components, ease of assembly, space and cost saving, are maintained. The device provides a high output current capability (up to 3.5A) very low harmonic and cross-over distortion.

Completely safe operation is guaranteed due to protection against DC and AC short circuit between all pins and ground, thermal over-range, load dump voltage surge up to 40V and fortuitous open ground.

Water Softener Circuit

That circuit is based at a technique to remove or neutralize the salt in water, and protect the pipes at home as well as the washing machines or our selves from salt. Its called water softener and its automated circuit using two 555 timers. The cost of parts is nearly 10$ and its very easy to build it.







HI-FI CAR-RADIO Amplifier

Constraints of implementing high power solutions are the power dissipation and the size of the power supply. These are both due to the low efficiency of conventional AB class amplifier approaches.


Here above is described a circuit proposal for a high efficiency amplifier which can be adopted for both HI-FI and CAR-RADIO applications. The TDA7294 is a monolithic MOS power amplifier which can be operated at 80V supply voltage (100V with no signal applied) while delivering output currents up to ±10 A.

This allows the use of this device as a very high power amplifier (up to 180W as peak power with T.H.D.=10 % and Rl = 4 Ohm); the only drawback is the power dissipation, hardly manageable in the above power range. Figure 20 shows the power dissipation versus output power curve for a class AB amplifier, compared with a high efficiency one. In order to dimension the heatsink (and the power supply), a generally used average output power value is one tenth of the maximum output power at T.H.D.=10 %.

From figure below, where the maximum power is around 200 W, we get an average of 20 W, in this condition, for a class AB amplifier the average power dissipation is equal to 65 W. The typical junction-to-case thermal resistance of the TDA7294 is 1 oC/W (max= 1.5 oC/W). To avoid that, in worst case conditions, the chip temperature exceedes 150 oC, the thermal resistance of the heatsink must be 0.038 oC/W (@ max ambient temperature of 50 oC). As the above value is pratically unreachable; a high efficiency system is needed in those cases where the continuous RMS output power is higher than 50-60 W.

The TDA7294 was designed to work also in higher efficiency way. For this reason there are four power supply pins: two intended for the signal part and two for the power part. T1 and T2 are two power transistors that only operate when the output power reaches a certain threshold (e.g. 20 W). If the output power increases, these transistors are switched on during the portion of the signal where more output voltage swing is needed, thus ”bootstrapping” the power supply pins (#13 and #15). The current generators formed by T4, T7, zener diodes Z1,Z2 and resistors R7, R8 define the minimum drop across the power MOS transistors of the TDA7294. L1, L2, L3 and the snubbers C9, R1 and C10, R2 stabilize the loops formed by the ”bootstrap” circuits and the output stage of the TDA7294.


In figures above, the performances of the system in terms of distortion and output power at various frequencies (measured on PCB shown in fig. 19) are displayed. The output power that the TDA7294 in high efficiency application is able to supply at
Vs = +40V/+20V/-20V/ -40V; f =1 KHz is:
- Pout = 150 W@ T.H.D.=10 % with Rl= 4 Ohm
- Pout = 120 W@ ” = 1% ” ” ”
- Pout = 100 W@ ” =10% with Rl= 8 Ohm
- Pout = 80 W @ ” = 1% ” ” ”

Results from efficiency measurements (4 and 8 Ohm loads, Vs = ±40V) are shown by figures 23
and 24. We have 3 curves: total power dissipation, power dissipation of the TDA7294 and power dissipation of the darlingtons. By considering again a maximum average output power (music signal) of 20W, in case of the high efficiency application, the thermal resistance value needed from the heatsink is 2.2oC/W (Vs =±40 V and Rl= 4 Ohm).
All components (TDA7294 and power transistors T1 and T2) can be placed on a 1.5oC/W heatsink, with the power darlingtons electrically insulated from the heatsink. Since the total power dissipation is less than that of a usual class AB amplifier, additional cost savings can be obtained while optimizing the power supply, even with a high headroom.

Wednesday, April 8, 2009

Automatic Room Lights

n ordinary automatic room power control circuit has only one light sensor. So when a person enters the room it gets one pulse and the lights come ‘on.’ When the person goes out it gets another pulse and the lights go ‘off.’ But what happens when two persons enter the room, one after the other? It gets two pulses and the lights remain in ‘off’ state. The circuit described here overcomes the above-mentioned problem. It has a small memory which enables it to automatically switch ‘on’ and switch ‘off’ the lights in a desired fashion. The circuit uses two LDRs which are placed one after another (separated by a distance of say half a metre) so that they may separately sense a person going into the room or coming out of the room. Outputs of the two LDR sensors, after processing, are used in conjunction with a bicolour LED in such a fashion that when a person gets into the room it emits green light and when a person goes out of the room it emits red light, and vice versa.



These outputs are simultaneously applied to two counters. One of the counters will count as +1, +2, +3 etc when persons are getting into the room and the other will count as -1, -2, -3 etc when persons are getting out of the room. These counters make use of Johnson decade counter CD4017 ICs. The next stage comprises two logic ICs which can combine the outputs of the two counters and determine if there is any person still left in the room or not. Since in the circuit LDRs have been used, care should be taken to protect them from ambient light. If desired, one may use readily available IR sensor modules to replace the LDRs.

The sensors are installed in such a way that when a person enters or leaves the room, he intercepts the light falling on them sequentially—one after the other. When a person enters the room, first he would obstruct the light falling on LDR1, followed by that falling on LDR2. When a person leaves the room it will be the other way round. In the normal case light keeps falling on both the LDRs, and as such their resistance is low (about 5 kilo-ohms). As a result, pin 2 of both timers (IC1 and IC2), which have been configured as monostable flip-flops, are held near the supply voltage (+9V).

When the light falling on the LDRs is obstructed, their resistance becomes very high and pin 2 voltages drop to near ground potential, thereby triggering the flip-flops. Capacitors across pin 2 and ground have been added to avoid false triggering due to electrical noise. When a person enters the room, LDR1 is triggered first and it results in triggering of monostable IC1.

The short output pulse immediately charges up capacitor C5, forward biasing transistor pair T1-T2. But at this instant the collectors of transistors T1 and T2 are in high impedance state as IC2 pin 3 is at low potential and diode D4 is not conducting. But when the same person passes LDR2, IC2 monostable flip-flop is triggered. Its pin 3 goes high and this potential is coupled to transistor pair T1-T2 via diode D4. As a result transistor pair T1-T2 conducts because capacitor C5 retains the charge for some time as its discharge time is controlled by resistor R5 (and R7 to an extent).

Thus green LED portion of bi-colour LED is lit momentarily. The same output is also coupled to IC3 for which it acts as a clock. With entry of each person IC3 output (high state) keeps advancing. At this stage transistor pair T3-T4 cannot conduct because output pin 3 of IC1 is no longer positive as its output pulse duration is quite short and hence transistor collectors are in high impedance state. When persons leave the room, LDR2 is triggered first followed by LDR1.

Since the bottom half portion of circuit is identical to top half, this time with the departure of each person red portion of bi-colour LED is lit momentarily and output of IC4 advances in the same fashion as in case of IC3. The outputs of IC3 and those of IC4 (after inversion by inverter gates N1 through N4) are ANDed by AND gates (A1 through A4) are then wire ORed (using diodes D5 through D8). The net effect is that when persons are entering, the output of at least one of the AND gates is high, causing transistor T5 to conduct and energise relay RL1.

The bulb connected to the supply via N/O contact of relay RL1 also lights up. When persons are leaving the room, and till all the persons who entered the room have left, the wired OR output continues to remain high, i.e. the bulb continues to remains ‘on,’ until all persons who entered the room have left. The maximum number of persons that this circuit can handle is limited to four since on receipt of fifth clock pulse the counters are reset.

The capacity of the circuit can be easily extended for up to nine persons by removing the connection of pin 1 from reset pin (15) and utilising Q1 to Q9 outputs of CD4017 counters. Additional inverters, AND gates and diodes will, however, be required.

Running Message Display

Light emitting diodes are advan- tageous due to their smaller size, low current consumption and catchy colours they emit. Here is a running message display circuit wherein the letters formed by LED arrangement light up progressively. Once all the letters of the message have been lit up, the circuit gets reset. The circuit is built around Johnson decade counter CD4017BC (IC2). One of the IC CD4017BE’s features is its provision of ten fully decoded outputs, making the IC ideal for use in a whole range of sequencing operations. In the circuit only one of the outputs remains high and the other outputs switch to high state successively on the arrival of each clock pulse.



The timer NE555 (IC1) is wired as a 1Hz astable multivibrator which clocks the IC2 for sequencing operations. On reset, output pin 3 goes high and drives transistor T7 to ‘on’ state. The output of transistor T7 is connected to letter ‘W’ of the LED word array (all LEDs of letter array are connected in parallel) and thus letter ‘W’ is illuminated. On arrival of first clock pulse, pin 3 goes low and pin 2 goes high. Transistor T6 conducts and letter ‘E’ lights up. The preceding letter ‘W’ also remains lighted because of forward biasing of transistor T7 via diode D21. In a similar fashion, on the arrival of each successive pulse, the other letters of the display are also illuminated and finally the complete word becomes visible. On the following clock pulse, pin 6 goes to logic 1 and resets the circuit, and the sequence repeats itself. The frequency of sequencing operations is controlled with the help of potmeter VR1.

The display can be fixed on a veroboard of suitable size and connected to ground of a common supply (of 6V to 9V) while the anodes of LEDs are to be connected to emitters of transistors T1 through T7 as shown in the circuit. The above circuit is very versatile and can be wired with a large number of LEDs to make an LED fashion jewellery of any design. With two circuits connected in a similar fashion, multiplexing of LEDs can be done to give a moving display effect.

Electronic Scoring Game

You can play this game alone or with your friends. The circuit comprises a timer IC, two decade counters and a display driver along with a 7-segment display.

The game is simple. As stated above, it is a scoring game and the competitor who scores 100 points rapidly (in short steps) is the winner. For scoring, one has the option of pressing either switch S2 or S3. Switch S2, when pressed, makes the counter count in the forward direction, while switch S3 helps to count downwards. Before starting a fresh game, and for that matter even a fresh move, you must press switch S1 to reset the circuit. Thereafter, press any of the two switches, i.e. S2 or S3.

On pressing switch S2 or S3, the counter’s BCD outputs change very rapidly and when you release the switch, the last number remains latched at the output of IC2. The latched BCD number is input to BCD to 7-segment decoder/driver IC3 which drives a common-anode display DIS1. However, you can read this number only when you press switch S4. The sequence of operations for playing the game between, say two players ‘X’ and ‘Y’, is summarised below:

Player ‘X’ starts by momentary pressing of reset switch S1 followed by pressing and releasing of either switch S2 or S3. Thereafter he presses switch S4 to read the display (score) and notes down this number (say X1) manually.
Player ‘Y’ also starts by momentary pressing of switch S1 followed by pressing of switch S2 or S3 and then notes down his score (say Y1), after pressing switch S4, exactly in the same fashion as done by the first player.
Player ‘X’ again presses switch S1 and repeats the steps shown in step 1 above and notes down his new score (say, X2). He adds up this score to his previous score. The same procedure is repeated by player ‘Y’ in his turn.
The game carries on until the score attained by one of the two players totals up to or exceeds 100, to be declared as the winner.

Several players can participate in this game, with each getting a chance to score during his own turn. The assembly can be done using a multipurpose board. Fix the display (LEDs and 7-segment display) on top of the cabinet along with the three switches. The supply voltage for the circuit is 5V

Tuesday, April 7, 2009

Wiper Speed Control

A continuously working wiper in a car may prove to be a nuisance, especially when it is not raining heavily. By using the circuit described here one can vary sweeping rate of the wiper from once a second to once in ten seconds. The circuit comprises two timer NE555 ICs, one CD4017 decade counter, one TIP32 driver transistor, a 2N3055 power transistor (or TIP3055) and a few other discrete components.


Timer IC1 is configured as a mono- stable multivibrator which produces a pulse when one presses switch S1 momentarily. This pulse acts as a clock pulse for the decade counter (IC2) which advances by one count on each successive clock pulse or the push of switch S1. Ten presets (VR1 through VR10), set for different values by trial and error, are used at the ten outputs of IC2.

But since only one output of IC2 is high at a time, only one preset (at selected output) effectively comes in series with timing resistors R4 and R5 connected in the circuit of timer IC3 which functions in astable mode. As presets VR1 through VR10 are set for different values, different time periods (or frequencies) for astable multivibrator IC3 can be selected.


The output of IC3 is applied to pnp driver transistor T1 (TIP32) for driving the final power transistor T2 (2N3055) which in turn drives the wiper motor at the selected sweep speed. The power supply for the wiper motor as well as the circuit is tapped from the vehicle’s battery itself. The duration of monostable multivibrator IC1 is set for a nearly one second period.

Ultrasonic Pest Repellant

This electronic circuit is an ultrasonic pest repellant are repelled by variabble ultrasonic frequency in the range of 30 kHz to 50 kHz. Thus to increase the effectiveness, frequency of ultrasonic oscillator has to be continuously varied between certain limits.


By using this circuit design, frequency of emission of ultrasonic sound is continuously varied step-by-step automatically. Here five steps of variation are used but the same can be extended up to 10 steps, if desired. For each clock pulse output from op-amp IC1 CA3130 (which is wired here as a low-frequency square wave oscillator), the logic 1 output of IC2 CD4017 (which is a well-known decade counter) shifts from Q0 to Q4 (or Q0 to Q9).

Five presets VR2 through VR6 (one each connected at Q0 to Q4 output pins) are set for different values and connected to pin 7 of IC3 (NE555) electronically. VR1 is used to change clock pulse rate. IC3 is wired as an astable multivibrator operating at a frequency of nearly 80 kHz. Its output is not symmetrical. IC4 is CD4013, a D-type flip-flop which delivers symmetrical 40kHz signals at its Q and Q outputs which are amplified in push-pull mode by transistors T1, T2, T3 and T4 to drive a low-cost, high-frequency piezo tweeter.

For frequency adjustments, you may use an oscilloscope. It can be done by trial and error also if you do not have an oscilloscope. This pest repeller would prove to be much more effective than those published earlier because here ultrasonic frequency is automatically changed to cover different pests and the power output is also sufficiently high. If you want low-power output in 30-50 kHz ultrasonic frequency range then the crystal transducer may be directly connected across Q and Q outputs of IC4 (transistor amplifier is not necessary).

Power Supply Circuit 12-15 Volt 20A

Output voltage of the power supply circuit is adjustable from fine potensiometer from 12V to 15v. It is suitable for all 12V power supply devices, or devices which are normally connected to a 12V battery or a vehicle with a 12V power supply system. This tension is usually 13.8 V.



For above reason, The Power Supply is also set to this tension, all right, however, any voltage from 12V to 14V. In this case, the tension is set somewhere around 13.6 V. To provide tension resistance in addition to voltage regulator 78S12. Instead potentiometer 100R inserted resistor 56R.



The scheme of the power supply is simple, but it is partly taken from some of the schemes taken up in the past. The material used is easily obtainable in electronic component shops, and this was the condition when I started to design this power supply.




ICL7107 Digital LED Voltmeter

This circuit is a digital voltmeter with LED display. It's ideal to use for measuring the output voltage of your DC power supply. It includes a 3.5-digit LED display with a negative voltage indicator. It measures DC voltages from 0 to 199.9V with a resolution of 0.1V. The voltmeter is based on single ICL7107 chip and may be fitted on a small 3cm x 7cm printed circuit board. The circuit should be supplied with a 5V voltage supply and consumes only around 25mA.



The use of 7805 5V voltage regulator is highly recommended to prevent the damage of ICL7107, 555 ICs and to extend the operating voltages.


Parts list of The Digital LED Voltmeter:


  • R1 = 8K2 R1 = 8K2
  • R2 = 47K / 470K R2 = 47k / 470K
  • R3 = 100K R3 = 100K
  • R4 = 2K R4 = 2K
  • R5, R6 = 47K R5, R6 = 47k
  • R7 = 0R / 4K7 R7 = 0R / 4K7
  • R8 = 560R R8 = 560R
  • C1,C5, C6, C8, C9 = 100n C1, C5, C6, C8, C9 = 100n
  • C2 = 470n / 47n C2 = 470n / 47n
  • C3 = 220n C3 = 220n
  • C4 = 100p C4 = 100p
  • C7 = 10-22u C7 = 10-22U
  • D1, D2 = 1N4148 D1, D2 = 1N4148
  • IC1 = ICL7107 IC1 = ICL7107
  • IC2 = NE555 IC2 = NE555
  • OPTO = CA 10 pin FTA = CA 10 pin
The digital LED voltmeter can also be configured to measure different voltage ranges and display higher voltage resolution.



Battery Low Voltage Beeper

This electronic circuit is an alarm circuit for low battery condition. It provides an audible and visual low voltage warning for 12V battery powered devices. When the battery voltage is above the set point (typically 11V), the circuit is idle. If the battery voltage should fall below the set point, the LED will light and the speaker will emit a periodic beeping sound to warn of the impending loss of power. The circuit was designed for monitoring solar systems, but it could also be useful for automotive and other 12V applications.



How it works

U2 provides a 5V regulated voltage reference. U1 is wired as a comparator, it compares the fixed 5V regulated voltage to the voltage on the wiper of VR1, that is proportional to the 12V supply. When the supply drops below the set point, the output of U1 goes low, turning on Q1 and powering the beeper and the LED.




The beeper consists of U4, a tone generator, and U3, a low duty cycle pulse generator. The tone can be changed by adjusting R7, the beep rate can be changed by adjusting R5. A small amount of hysteresis is provided by R1 and the current through LED1 and the beeper, this separates the on and off points for the circuit.

U2 provides a 5V regulated voltage reference. U1 is wired as a comparator, it compares the fixed 5V regulated voltage to the voltage on the wiper of VR1, that is proportional to the 12V supply. When the supply drops below the set point, the output of U1 goes low, turning on Q1 and powering the beeper and the LED.

The beeper consists of U4, a tone generator, and U3, a low duty cycle pulse generator. The tone can be changed by adjusting R7, the beep rate can be changed by adjusting R5. A small amount of hysteresis is provided by R1 and the current through LED1 and the beeper, this separates the on and off points for the circuit.

Use of Battery Low Voltage Beeper

Connect the circuit to the 12V source that you wish to monitor. Turn S1 on, if the battery voltage is above the set point, nothing should happen.

As the battery voltage drops below the set point, the LED will light and a periodic beeping will come from the speaker. If the beeping becomes annoying, turn off S1. Be sure to charge the battery soon, excessive discharging will shorten the life of most rechargeable batteries.

Car Anti-Theft Wireless Alarm

This alarm circuit is an anti- theft wireless alarm can be used with any vehicle having 6- to 12-volt DC supply system. The mini VHF FM radio-controlled, FM transmitter is fitted in the vehicle at night when it is parked in the car porch or car park.



The receiver unit of the wireless alarm uses an CXA1019, a single IC-based FM radio module, which is freely available in the market at reasonable rate, is kept inside. Receiver is tuned to the transmitter's frequency. When the transmitter is on and the signals are being received by FM radio receiver, no hissing noise is available at the output of receiver. Thus transis- tor T2 (BC548) does not conduct. This results in the relay driver transistor T3 getting its forward base bias via 10k resistor R5 and the relay gets energised.


When an intruder tries to drive the car and takes it a few metres away from the car porch, the radio link betw- een the car (transmitter) and alarm (receiver) is broken. As a result FM radio module gene-rates hissing noise. Hissing AC signals are coupled to relay switching circ- uit via audio transformer. These AC signals are rectified and filtered by diode D1 and capacitor C8, and the resulting positive DC voltage provides a forward bias to transistor T2. Thus transistor T2 conducts, and it pulls the base of relay driver transistor T3 to ground level. The relay thus gets de-activated and the alarm connected via N/C contacts of relay is switched on.

If, by chance, the intruder finds out about the wireless alarm and disconnects the transmitter from battery, still remote alarm remains activated because in the absence of signal, the receiver continues to produce hissing noise at its output. So the burglar alarm is fool-proof and highly reliable. (Ed: You may have some problem catching the thief, though, if he decides to run away with your vehicle_in spite of the alarm!)

2-Way Electronic Crossover Network

The electronic crossover featured here is an 18dB / octave unit, and has the crossover frequency centred on 300Hz. The frequency may be changed by increasing (or decreasing) resistor / capacitor values.

Increasing capacitance or resistance - Reduces frequency
  • Doubling the capacitance or resistance halves the frequency
Reducing capacitance or resistance - Increases frequency
  • Halving the capacitance or resistance doubles the frequency

The values of resistance and capacitance (indicated with a * in the circuit diagram) in the filter are critical, and close tolerance components are mandatory. If you cannot obtain close tolerance capacitors, use a capacitance meter to select values within 5% of the indicated value. Use only 1% metal film resistors throughout. The 1uF coupling caps are not critical, and standard tolerance is Ok.


If the crossover frequency is changed, it is critical that the ratios of capacitor and resistor values are not varied. For example, if you wanted to halve the frequency, the resistors would become 22k and 102k (100k is only just acceptable. If the ratios are changed, the filter damping is also changed, and the behaviour at the crossover point will be unpredictable (causing a dip or peak in the frequency response).

The values you change to alter the crossover frequency are indicated with a * in the circuit diagram

Do not change the 10k resistors - they set the damping of the filter and strange happenings will befall s/he who fiddles indiscriminately.

the NE5532 Dual op-amp is used. This circuit can be operated from the same power supply as the Audio Preamp, featured elsewhere on these pages. Other dual opamps may also be used, depending on your preference.
The input is buffered by U1a (the second channel can use the other half of the op-amp), and fed to the two filter networks. Each filter is a 3rd order section, and has a gain of 2. The output of each section is fed (via a 1uF polyester capacitor) to the level control and output buffer stage.
In use, the output of the preamplifier is fed to the input of the crossover network, and the outputs are fed to their respective amplifiers. For more information on bi-amping, refer to the article "Bi-Amplification - Not quite magic (but close)" on these pages.

Be careful when adjusting the level controls, since it is easy to create a mismatch in levels between the amplifiers. I suggest that the controls be mounted on the rear panel, with their shafts cut off really short, and a slot cut into the end with a hacksaw. Once the adjustment is made, it should not require further changes in use. Make sure that the power amplifier volume controls (if fitted) are turned fully up, and try to set the crossover controls so somewhere between midway and 75%. This ensures plenty of scope for getting the levels right, and will ensure that the preamp settings are not radically different from their "pre-biamp" days.

Classic PIC Programmer

PICs are small microprocessors containing RAM, ROM, and some peripherals. Almost no other parts are required to make a complete “embedded system”. They are readily available and well supported by the manufacturer, third party developers, and most importantly, users. This has led to their immense popularity.


Assembly:

The PC board design is fairly straightforward and can be made by laser printing to special paper or a page from TIME magazine, then ironing the image onto copper-clad board, then etching with ferric chloride. There are a few jumper wires. The power source needs to be at least 15 volts. A 12 volt DC adapter usually produces about 17 volts, so that's a good choice. Two 9-volt batteries in series will work too. Solder directly to the PC board or use a connector that mates with your power source. Pay attention to the direction of the voltage regulators because the plastic regulators are backward from the tab type. Substitute Japanese or European generic equivalents for the transistors and diodes, but remember that the pinouts will be different. A right angle PC mount DB-25M connector is specified, but a conventional solder-cup DB-25M connector works, see the picture how I did it.


Operation:

The programmer connects to the parallel printer port of your computer and requires external power. If you want to program a PIC you'll have a hex file created by your assembler or created by someone else(see my propeller clock). You will also need to drive the programmer with some software. Here are programs that run under DOS and Windows. Linux software for Intel-based computers is available elsewhere. Macintoshes do not have parallel ports and can't use this programmer. Do not insert the PIC to be programmed until you have power applied and have run the software, and the programming LED is not lit. The DOS software requires command line switches for fuse settings(unless in the chip's hex file)and also the environmental variable "set ppsetup=3" to be typed before running the program. The Windows software requires the driver "dtait.drv" to be in the \windows\system directory and also the line "PINAPI=DTAIT.DRV" added to the system.ini file. Tell the software you have 7407 chip and PNP transistors. These details are explained in the text files included with the software.

The Files:

DOS software by David Tait "pic84v05.zip".
Windows software v1.03 by Silicon Studio "picser.zip".

Programming newer PICs:
  • The whole “F” series can be programmed. You need to use newer software, like this cool software. The PIC16F627 and PIC16F628 are 18 pin devices and fit right in the socket, but you must make a ground connection to pin 10 to prevent LVP programming, a new feature this programmer does not use. Some people suggest using a 10K resistor to ground, if you are doing in-circuit programming that probably makes sense. Programming the bigger PICs, including the PIC16F872 through 16F877 requires fitting the correct(28 or 40 pin) socket and wiring the pins to the corresponding function. Remember to ground the LVP pin on these, too. The bigger PICs also have extra power and ground pins. These must all be used.
  • You must select the port your computer is using(usually 0378) and the type of programmer (P16PRO) and the type of buffers the programmer uses (non-inverting). The software is beta, but I have tested it and can testify it working on the 16F84 and 16F628 I tried. It only programs locations used in the hex file, so it is very fast. If your program is 250 bytes, only 250 bytes get programmed, but when I used my PicstartPlus to verify the chips I tried, it would show a verify error unless I blanked the chip first, although the chip functioned fine. Leaving those unused areas in the previously programmed state shouldn’t be a problem.
  • Propic2 keeps the power to the chip on while idle. This can be useful for “burn and crash” in-circuit programming. You’ll see the LED is lit. I don’t like to insert or remove the PIC when power is present, so I pull the power cord before I insert or remove the PIC.
  • propic2 software, in case the above link is broken.