Electronics Circuit Application

LOW-COST HEARING AID

2009-06-13T20:20:00.000-07:00

Commercially available hearing aids are quite costly. Here is an inexpensive hearing aid circuit that uses just four transistors and a few passive components. On moving power switch S to ‘on’ position, the condenser microphone detects the sound signal, which is amplified by transistors T1 and T2. Now the amplified signal passes through coupling capacitor C3 to the base of transistor T3. The signal is further amplified by pnp transistor T4 to drive a low impedance earphone. bg8j9xp2ym

Capacitors C4 and C5 are the power supply decoupling capacitors. The circuit can be easily assembled on a small, general-purpose PCB or a Vero board. It operates off a 3V DC supply. For this, you may use two small 1.5V cells. Keep switch S to ‘off’ state when the circuit is not in use. To increase the sensitivity of the condenser microphone, house it inside a small tube. This circuit costs around Rs 65.

Magnetic Levitation

2009-06-12T20:42:00.000-07:00

How Levitation Works

If you hold two permanent magnets close together, you see that one of them will jump strongly toward (or away) from the other. In 1842, Samuel Earnshaw expressed the perversity of inanimate magnetic objects in his theorem. It explains this frustrating behavior will always prevent you from suspending one permanent magnet above or below another, no matter how one arranges the two magnets. However, an active control circuit can get around this problem by rapidly adjusting the magnet's strength.

Click this image to see closeup of antigravity relay (38K) The general principle is straight forward: An electromagnet pulls a ball upward while a light beam measures the exact position of the ball's top edge. The magnet's lifting force is adjusted according to position.

As less light is detected, the circuit reduces the electromagnet's current. With less current, the lifting effect is weaker and the ball can move downward until the light beam is less blocked. Voila! The ball stays centered on the detector! It is a small distance across the photodetector, perhaps a millimeter or two, but this is sufficient to measure small changes in position. Of course, if the ball is removed the coil runs at full power. And conversely, if the light beam is blocked the coil is turned completely off.

Click here to see image of breadboard circuit (54K) This device uses two photodetectors: the "signal" detector looks for an interruption in the light beam, and the "reference" detector measures the background light. The circuit subtracts one signal from the other to determine the ball's position. The use of two detectors is my small contribution to advance the art of levitation. This design automatically compensates for changes in ambient light, and eliminates a manually

Magnetic Levitation Parts List

Resistors

Resistors listed in order by value are 1/4-watt, 5% unless otherwise indicated.

300 ohms R11
500 ohms R2
1,000 ohms R1, R12, R13, R14
1,500 ohms R10
10,000 ohms R4
11,000 ohms R6
22,000 ohms R8
56,000 ohms R3
100,000 ohms R5
150,000 ohms R7
370,000 ohms R9
50K linear taper VR1 (and VR2 opt.)

Capacitors

C1,C2 47 uF electrolytic
C3 0.1 uF ceramic or tantalum (must not be electrolytic)

Semiconductors

Q1,Q2 OP505A infrared photo detector, or equivalent
Q3 2N3055 NPN power transistor
LED1,2,4 Red light-emitting diode
LED 3 Infrared LED emitter
IC1-4 LM741 op amp, Radio Shack 276-007
D1 1N4001 (or 1N4004) silicon diode, 50v (or more) peak inverse voltage

Miscellaneous

+/- 15 vdc power supply, 1 amp
9 vdc power supply, 1 amp
Breadboard wiring pad (or printed circuit board by Amadeus)
18-ga stranded wire for power
Solid hook-up wire
24-ga (or thicker) magnet wire for lifting coil
6-terminal barrier strip (2 ea.)
Wood for base and frame
Alternatives for the LM741 Op-Amp

I chose the LM741 op-amp out of nostalgia and convenience. It was an extremely successful and common op-amp about twenty years ago.

There are lots of modern choices for dual- and quad-package op-amps. By using a package with multiple op-amps, you can reduce the number of parts and lower the cost. For example, you could use a single quad-package op-amp instead of four separate 741s. This would allow a very small printed circuit board to contain all the electronics!

Electronic Stethoscope

2009-06-12T20:32:00.000-07:00

Stethoscopes are not only useful for doctors, but home mechanics, exterminators, spying and any number of other uses. Standard stethoscopes provide no amplification which limits their use. This circuit uses op-amps to greatly amplify a standard stethoscope, and includes a low pass filter to remove background noise.

Parts

Part Total Qty. Description

R1 ---------------1 ----------- 10K 1/4W Resistor
R2 ---------------1 ----------- 2.2K 1/4W Resistor
R4 ---------------1------------ 47K 1/4W Resistor
R5, R6, R7 -------3------------ 33K 1/4W Resistor
R8 ---------------1 ----------- 56K 1/4W Resistor
R10 --------------1 ----------- 4.7K 1/4W Resistor
R11 --------------1 ------------ 2.2K to 10K Audio Taper Pot
R12 --------------1------------ 330K 1/4W Resistor
R13, R15, R16---- 3------------ 1K 1/4W Resistor
R14 --------------1 ----------- 3.9 Ohm 1/4W Resistor
C1, C8 -----------2 ---------- 470uF 16V Electrolytic Capacitor
C2 ---------------1-----------4.7uF 16V Electrolytic Capacitor
C3, C4 -----------2----------- 0.047uF 50V Metalized Plastic Film Capacitor
C5 ---------------1----------- 0.1uF 50V Ceramic Disc Capacitor
C6, C7 -----------2----------- 1000uF 16V Electrolytic Capacitor
U1 ---------------1 ----------- TL072 Low Noise Dual Op-Amp
U4 ---------------1----------- 741 Op-Amp
U5 ---------------1----------- LM386 Audio Power Amp
MIC --------------1 -----------Two Wire Electret Microphone
J1 ----------------1 -----------1/8" Stereo Headphone Jack
Batt1, Batt2 ------2----------- 9V Alkaline Battery
LED --------------1 ----------- Red/Green Dual Colour Two Wire LED
SW ---------------1 ----------- DPST Switch
MISC -------------1 ----------- Stethoscope head or jar lid, rubber sleeve for microphone, board, wire, battery clips, knob for R11

Notes

MIC is an assembly made out of a stethoscope head and electret mic. Cut the head off the stethoscope and use a small piece of rubber tube to join the nipple on the head to the mic.
Be careful with the volume, as excess noise levels may damage your ears.
R11 is the volume control.
The circuit marked as optional is not required for the main circuit to function. The optional circuit blinks an LED to the heartbeat as it is heard by the microphone. Even if the optional circuit is not included, sound will still be heard via the headphone jack.

Muscular Bio-Stimulator

2009-06-12T20:30:00.000-07:00

Particularly suitable for cellulite treatment
3V battery supply, portable set

Device purpose:

This is a small, portable set, designed for those aiming at look improvement. The Bio-Stimulator provides muscles' stimulation and invigoration but, mainly, it could be an aid in removing cellulite.

Tape the electrodes to the skin at both ends of the chosen muscle and rotate P1 knob slowly until a light itch sensation is perceived. Each session should last about 30 - 40 minutes.

Parts:

P1______________4K7 Linear Potentiometer
R1____________180K 1/4W Resistor
R2______________1K8 1/4W Resistor (see Notes)
R3______________2K2 1/4W Resistor
R4____________100R 1/4W Resistor
C1____________100nF 63V Polyester Capacitor
C2____________100µF 25V Electrolytic Capacitor
D1______________LED Red 5mm.
D2___________1N4007 1000V 1A Diode
Q1,Q2_________BC327 45V 800mA PNP Transistors
IC1____________7555 or TS555CN CMos Timer IC
T1_____________220V Primary, 12V Secondary 1.2VA Mains transformer (see Notes)
SW1____________SPST Switch (Ganged with P1)
B1_____________3V Battery (two 1.5V AA or AAA cells in series etc.)

Warning:

The use of this device is forbidden to Pace-Maker bearers and pregnant women.
Do not place the electrodes on cuts, wounds, injuries or varices.
Obviously we can't claim or prove any therapeutic effectiveness for this device.

Circuit operation:

IC1 generates 150µSec. pulses at about 80Hz frequency. Q1 acts as a buffer and Q2 inverts the polarity of the pulses and drives the Transformer. The amplitude of the output pulses is set by P1 and approximately displayed by the brightness of LED D1. D2 protects Q2 against high voltage peaks generated by T1 inductance during switching.

Notes:

T1 is a small mains transformer 220 to 12V @ 100 or 150mA. It must be reverse connected i.e. the 12V secondary winding across Q2 Collector and negative ground, and the 220V primary winding to output electrodes.
Output voltage is about 60V positive and 150V negative but output current is so small that there is no electric-shock danger.
In any case P1 should be operated by the "patient", starting with the knob fully counter-clockwise, then rotating it slowly clockwise until the LED starts to illuminate. Stop rotating the knob when a light itch sensation is perceived.
Best knob position is usually near the center of its range.
In some cases a greater pulse duration can be more effective in cellulite treatment. Try changing R2 to 5K6 or 10K maximum: stronger pulses will be easily perceived and the LED will shine more brightly.
Electrodes can be obtained by small metal plates connected to the output of the circuit via usual electric wire and can be taped to the skin. In some cases, moistening them with little water has proven useful.
SW1 should be ganged to P1 to avoid abrupt voltage peaks on the "patient's" body at switch-on, but a stand alone SPST switch will work quite well, provided you remember to set P1 knob fully counter-clockwise at switch-on.
Current drawing of this circuit is about 1mA @ 3V DC.
Some commercial sets have four, six or eight output electrodes. To obtain this you can retain the part of the circuit comprising IC1, R1, R2, C1, C2, SW1 and B1. Other parts in the diagram (i.e. P1, R3, R4, D1, D2, Q2 & T1) can be doubled, trebled or quadrupled. Added potentiometers and R3 series resistors must be wired in parallel and all connected across Emitter of Q1 and positive supply.
Commercial sets have frequently a built-in 30 minutes timer. For this purpose you can use the Timed Beeper the Bedside Lamp Timer or the Jogging Timer circuits available on this Website, adjusting the timing components to suit your needs.

Automated Crib Lights

2009-06-12T20:12:00.000-07:00

Device purpose:

This circuit is intended to drive the various lights decorating the crib prepared during Christmas season at many homes in Latin Countries, especially for children delight, in order to obtain realistic light-effects.

Features:

Alternating day and night with lights gradually dimming from full-on to full-off and the opposite.
Slow turn on of model-houses interior as night approaches, and slow turn off as sun rises, with presettable intensity, thus imitating candles' light for a more realistic effect.
Flickering ever-running circuit driving bulbs for fires, firesides, lanterns effects etc.
Total cycle duration: 2 minutes. Day duration: 1 minute, 15 seconds. Night duration: 45 seconds. (All values are approximate).

Parts:

R1___________150K 1/4W Resistor
R2,R9,R14_____22K 1/4W Resistors
R3,R11_______220K 1/4W Resistors
R4,R12________10K 1/4W Resistors
R5___________100K 1/2W Trimmer Cermet
R6,R7,R13,R15__1R 1/4W Resistors
R8____________33K 1/4W Resistor
R10__________270K 1/4W Resistor
R16___________47R 1/4W Resistor
C1,C4________100nF 63V Polyester Capacitors
C2,C6_________10µF 25V Electrolytic Capacitors
C3,C5________100µF 25V Electrolytic Capacitors
D1-D3_______1N4148 75V 150mA Diodes
IC1___________4060 14 stage ripple counter and oscillator IC
IC2__________LM324 Low power Quad Op-Amp IC
IC3__________78L12 12V 100mA Voltage regulator IC
Q1,Q3,Q5_____BC238 25V 100mA NPN Transistors
Q2,Q4,Q6_____BD681 100V 4A NPN Darlington Transistors
J1___________Miniature input socket,
suited for commercial plug-in variable voltage power supplies
J2-J5________Two ways output sockets

Load requirements:

Input J1 is connected to a commercial wall plug-in power supply transformer adapter with variable output settled to 12-15Vdc, and a required minimum output capability of 600mA @ 12V. Using a good number of bulbs the output capability must reach about 1.5A.
Output J2 is connected to a permanently-on 12V 1W blue bulb(s) for night effect.
Output J3 is connected to several 12V 2.2W bulbs in parallel for sunlight effect. Max. output current: 1.2A (i.e. 6-7 bulbs).
Output J4 is connected to several 12V 1W or 1/2W micro-bulbs in parallel for house-interiors lights. Max. output current: 600mA (i.e. 7-8 1W bulbs, doubling in number if 1/2W).
Output J5 is connected to one or several 12V 1W or 1/2W micro-bulbs in parallel for fires, firesides, lanterns effects etc. Max. output current: 600mA (bulbs total number same as above).
All outputs are current limited, and short-proof for a reasonable lapse of time.

Circuit operation:

IC1 oscillates at a frequency calculated to obtain a pin 2 level change around every minute. IC2A is then enabled to slowly charge and discharge C5 through R10 during a 2 minutes period. IC1 pin 9 drives D2, R3 & C4, generating a sawtooth for IC2C & IC2D comparators. IC2D comparing the voltage at pin 13 with the sawtooth, generates a squarewave with variable mark-space ratio driving the output darlington Q2 for daylight bulbs. IC2B changes its output at a threshold voltage settled by R8 & R9, activating IC2C & Q4 that act like IC2D & Q2 driving model-houses bulbs as evening approaches and turning them off at dawn. R11 & C6 provide slow turn on and off and R5 sets the basic brightness of these bulbs. IC1 pin 7 drives the output darlington Q6 for flickering fires bulbs and R16 prevents them to turn off completely for a more realistic effect. Q1, Q3, Q5 and associated Base resistors provide current limiting.

Notes:

Total period length can be varied changing C1 and/or R1 values.
Day-night ratio can be varied changing R10 value slightly.
Threshold voltage of turn on and off of model-houses lights can be varied changing slightly R8 and/or R9 values.
Turn on and off speed of model-houses lights can be varied changing R11 value.
Current limiting can be varied changing Q2, Q4 & Q6 Emitter resistors.
Heatsinks for Q2, Q4 & Q6 are needed if current limits are increased.

Cranial Electrotherapy Stimulator

2009-06-12T20:00:00.000-07:00

Current generated flows through clips placed on the earlobes Output current adjustable from 80 to 600 microAmperes

Device purpose:

Owing to the recent launching in Europe of Cranial Electrotherapy Stimulation (CES) portable sets, we have been "Electronically Stimulated" in designing a similar circuit for the sake of Hobbyists. CES is the most popular technique for electrically boosting brain power, and has long been prescribed by doctors, mainly in the USA, for therapeutic reasons, including the treatment of anxiety, depression, insomnia, and chemical dependency. CES units generate an adjustable current (80 to 600 microAmperes) that flows through clips placed on the earlobes. The waveform of this device is a 400 milliseconds positive pulse followed by a negative one of the same duration, then a pause of 1.2 seconds. The main frequency is 0.5 Hz, i.e. a double pulse every 2 seconds. Some people report that this kind of minute specialized electrical impulses contributes to achieve a relaxed state that leaves the mind alert.

Obviously we can't claim or prove any therapeutic effectiveness for this device, but if you are interested in trying it, the circuit is so cheap and so simple to build that an attempt can be made with quite no harm.

Parts:

R1_____________1M5 1/4W Resistor
R2____________15K 1/4W Resistor
R3___________100K Linear Potentiometer
R4_____________2K2 1/4W Resistor
C1___________330nF 63V Polyester Capacitor
C2___________100µF 25V Electrolytic Capacitor
D1_____________3mm. Red LED
IC1___________7555 or TS555CN CMos Timer IC
IC2___________4017 Decade counter with 10 decoded outputs IC
SW1___________SPST Slider Switch
B1______________9V PP3 Battery
Clip for PP3 Battery
Two Earclips with wires (see notes)

Circuit operation:

IC1 forms a narrow pulse, 2.5Hz oscillator feeding IC2. This chip generates the various timings for the output pulses. Output is taken at pins 2 & 3 to easily obtain negative going pulses also. Current output is limited to 600µA by R2 and can be regulated from 80 to 600µA by means of R3. The LED flashes every 2 seconds signaling proper operation and can also be used for setting purposes. It can be omitted together with R4, greatly increasing battery life.

Notes:

In order to obtain a more precise frequency setting take R1=1M2 and add a 500K trimmer in series with it.
In this case use a frequency meter to read 2.5Hz at pin 3 of IC1, or an oscilloscope to read 400msec pulses at pins 2, 3 or 10, adjusting the added trimmer.
A simpler setting can be made adjusting the trimmer to count exactly a LED flash every 2 seconds.
Earclips can be made with little plastic clips and cementing the end of the wire in a position suited to make good contact with earlobes.
Ultra-simple earclips can be made using a thin copper foil with rounded corners 4 cm. long and 1.5 cm. wide, soldering the wire end in the center, and then folding the foil in two parts holding the earlobes.
To ensure a better current transfer, this kind of devices usually has felt pads moistened with a conducting solution interposed between clips and skin.
Commercial sets have frequently a built-in timer. Timing sessions last usually 20 minutes to 1 hour. For this purpose you can use the Timed Beeper the Bedside Lamp Timer or the Jogging Timer circuits available on this website, adjusting the timing components in order to suit your needs.

Active Antenna HF/VHF/UHF, 3-3000MHz

2009-06-05T22:49:00.000-07:00

If you have a shortwave or high-frequency receiver or scanner that is struggling to capture signals with a short, whip antenna, and you'd like the kind of performance that a 60-foot 'longwire' antenna can provide but lack the space to put one up, consider building the AA-7 HF/VHF/UHF Active Antenna described in this article. The AA-7 is a relatively simple antenna that is designed to amplify signals from 3 to 3000 MegaHertz, including three recognized ranges: 3-30Mhz high-frequency (HF) signals; 3-300Mhz very-high frequency (VHF) signals; 300-3000MHz ultra-high (UHF) frequency signals. Those bands are typically occupied by shortwave, ham, government, and commercial radio signals.

Active Antennas:

In its simplest form, an active antenna uses a small whip antenna that feeds incoming RF to a pre-amplifier, whose output is then connected to the antenna input of a receiver. Unless specifically designed otherwise, all active antennas are intended for receive-only operation, and thus should not be used with transceivers; transmitting into an active antenna will probably destroy its active components. A well designed broadband active antenna consider field strength of the desired signal (measured in microvolts per meter of antenna length), atmospheric and other noise, diameter of the antenna, radiation resistance, and antenna reactance at various frequencies, plus the efficiency and noise figure of the amplifier circuit itself.

Circuit Description:

Figure shows the schematic diagram of the AA-7, which contains only two active elements; Q1 (an MFE201 N-Channel dual-gate MOSFET) and Q2 (a 2SC2570 NPN VHF silicon transistor). Those transistors provide the basis of two independent, switchable RF pre-amplifiers. Two double-pole double-throw (DPDT) switches play a major role in this operation of the AA-7. Switch S1 is used to select one of the two pre-amplifier circuits (either HF or VHF/UHF). Switch 2 is used to turn off the power to the circuit, while coupling the incoming RF directly to the input of the receiver. That gives the receiver non-amplified access to the auxiliary antenna jack, at J1, as well as the on-board telescoping whip antenna. With switch S2 in its power-on position, the input and output jacks are disconnected and B1 (a 9 volt battery) is connected to the circuit. With switch S1 in the position shown in the schematic, incoming RF is directed to the HF pre-amp circuit built around Q1 (an MFE201 N-Channel dual-gate MOSFET). The HF pre-amp operates with an exceptionally low noise level, and is ideal for copying weak CW and singe-side band signals. When S1 is switched to the other position, the captured signal is coupled to the VHF/UHF pre-amp built around Q2 (a 2SC2570 NPN VHF silicon transistor), which has excellent VHF through microwave characteristics. With the on-board whip antenna adjustable to resonance through much of the VHF-UHF region (length in feet = 234 divide by the frequency in MHz), the VHF/UHF mode is ideal for indoor and portable use with VHF scanners and other receivers. Either mode can be used when tuning 3-30 MHz HF signals. The VHF/UHF pre-amp offers higher gain than the HF pre-amp, but also has a higher noise level. You can easily choose either amplifier for copying any signal; of interest--just try both positions. The RF gain control (R5) can be used to trim the output of either amplifier.

Caution: The AA-7 is not intended for transmitting operation (be it Ham, Maritime, or CB); if it is used with a transceiver of any kind, make sure it is not possible to transmit by accidentally pressing a mike button or CW keyer. Transmitting RF into the AA-7 is likely to ruin one or both of the transistors in the circuit.

Parts List and other components:

Semiconductors:

Q1 = MFE201, SK3991, or NTE454. N-Channel, dual-gate MOSFET, TO-72 (see text)
Q2 = 2SC2570, NTE107. Silicon RF IF/Amp, NPN transistor (see text)
Note: If you use the NTE107 as a replacement, make sure to insert it correctly
into the pcb. The orientation is different than as shown on the parts layout
diagram. (e-c-b seen front view for NTE107). See this Data Sheet

Resistors:

All Resistors are 5%, 1/4-watt
R1 = 1 Mega Ohm
R2 = 220K
R3,R6 = 100K
R4 = 100 ohm
R5 = 10K potentiometer (pc mount)
Capacitors:

C1,C2,C5,C6 = 0.01uF, ceramic disc
C3 = 100pF ceramic disc
C4 = 4.7 to 10uF, 16WVDC, radial lead electrolytic

Additional Parts & Materials:

B1 = 9-volt alkaline battery
S1,S2 = DPDT PC mount pushbutton switch
J1,J2 = PC mount RCA jack
ANT1 = Telescoping whip antenna (screw mount)
MISC = PCB materials, enclosure, enclosure, battery holder and connector,
wire, solder, etc.

Motorcycle Battery Charger

2009-06-05T22:29:00.000-07:00

This 3A charger was originally designed to work with small batteries like those used in motorcycles. In principle it can be used to charge car batteries also but will take a lot longer.
The charger below charges a battery with a constant current to 14.1 volt. When this level is reached, the current charge drops automatically to a safer level (13.6V) and keeps charging at this slower rate untill the LED lights up indicating a fully charged battery. This project looks very much alike with the Gel cell II charger elsewhere posted in the 'Circuits' section. The difference is the IC, namely a LM1458 instead of a LM301A. Nice job Jan!

Description:

The LM350 is an adjustable voltage regulator and keeps the voltage between points C and B at 1.25 Volt. By adding a 1K resistor between point B and gnd (-) you can, as it were, lift up the output voltage. To accurately control the output voltage we add to this resistor, in series, a 2K adjustable 10-turn potentiometer. As soon as a battery is connected a current flow occurs, controlled by the right halve of the LM1458. The current through the 0.1 ohm resistor causes a voltage drop. This drop is compared with the voltage on the walker of 100-ohm pot. The moment this drop is greater than the one adjusted with the potmeter will cause the output of the LM1458 IC to go low and a small current starts to flow thru the diode and this in effect will reduce the current through the series resistors 1K + 2Kpot. The current is hereby stabilized.

The point between C and B is devided by three resistors; 2.2 ohm, 100 ohm pot, and the 150 ohm. 2.2 ohm and the 100 ohm potmeter are connected to the non-inverting input (+) of the LM1458 IC. The inverting input (-) is connected to the 0.1 ohm wire-wound resistor in series with the output. As long as the voltage drop, caused by the current-flow over this resistor is greater than the voltage drop over the 2.2 ohm resistor the output of the LM1458 will stay high and in turn block the BC558 transistor. But as soon as the charge current falls below a specific value the 1458 will go low and turn on this transistor which wich activate the LED. At the same time a small current will flow thru the 'Rx' resistor, which will cause that the output voltage of the charger switches to 13.6 Volt. This is a very safe output voltage, and does not cause overcharging to the battery and remains fully charged (trickle).

Rx should be an experimental value determined below; a mathematically calculation is possible but the exact value is determined by the tolerances of your specific components.

The voltage regulator LM350 has to dissipate a lot of energy so make sure to mount it on a large cooling fin. (e.i. 3.3°C/Watt) Diode 1N4001 over the input/output is necessary to prevent damage to the regulator in case the input voltage gets interrupted.

The LM350 can be substituted with a NTE970, and the BC558B with a NTE159 if you wish.
The adjustments for this charger are really simple and the only thing needed is digital multimeter. The LM1458 should NOT be in the socket while doing the first adjustment. When no battery is connected there is no current flow thru the 0.1 ohm resistor and therefore pulling the output low. So no IC yet in the socket. Do NOT connect a battery also. I know that is obvious to most of us, but some people... :-)

Okay, here we go:

Connect the multimeter (set for Volt DC) to the '+' and '-' battery output and adjust with the 2k trimpot the output voltage to 14.1 Volt.
Switch the power off. Discharge the capacitors (short them out with a piece of wire).
Now insert the LM1458 IC carefully (check no pins are bend underneath the chip).
Switch the power back on and make the resistor marked Rx such a value that the output voltage reads 13.6 volt exactly.
Switch the multimeter to 'Amp-dc'. Turn the 100-ohm trimpot all the way CCW. Connect the 'to-be-charged-battery' (e.i. NOT a fully charged battery) and turn back the trimpot untill the current load is 0.1 X the battery capacity (max 3A). Example: A 16Amp battery adjusting to 1.6A. If you don't have an Amp meter on your multimeter you can use the 2-volt setting on your meter and connect it over the 0.1 ohm resistor. The current is volt devided by 0.1, so for 3A the meter should read 0.3 volt.

That's it. To get the Rx value you could also use a trimpot until you get the 13.6volt and then read the ohm's value of the trimpot and replace with a resistor. In my opinion this resistor should be a metalfilm type at 1 or 2% tolerance.

The Technical bits:

For those of you interested in how the value of essential components was calculated, read on. You may be able to design your own charger for use with a different current or voltage (like 6-volt).
Calculations origin from the voltage between points C and B of the LM350 regulator. When a resistor is connected between these two points, enough current starts to flow that the voltage over this resistor measures 1.25 volt. In our case, the resistor total is 2.2 + 100 + 150 =252.2 ohm. Because we deal with very small currents the calculations are performed in milliamps and the calculations of resistance in Kilo-Ohms. Thus, the current thru this resistor is 1.25 / 0.2522 = 4.9564 mA. The same current also flows thru the 1K & 2K series resistors. We want the output voltage to be 14.1 volt, meaning the voltage drop over these series resistors must be 14.1 - 1.25 = 12.85 Volt.

The total resistance value thus must be 12.85 / 4.9564 = 2.5926 Ohms. To enable us to adjust it to this value, one of the resistors is chosen as a 10-turn trimpot (trimmer potentiometer). Together with the 1K in series (making it a total of 3K)we can adjust it to this correct value.

The Rx value is calculated this way; In this scenario we like to have a output voltage of 13.6 volt, in other words, the voltage on the connection point between the 1K/2Kpot should be 13.6 - 1.25 = 12.35 volt. This means that the current thru the 'voltage-divider' will be 12.35 / 2.5926 = 4.7635 mA and the leftover current should be 4.9564 - 4.7635 = 0.1929 mA thru Rx and also cause a voltage drop of 12.35 - 2.78 = 9.57 volt. Measuring this calculated value at the base of the BC558 transistor was 2.78 volt after the output of the LM1458 had become low. With the current of 0.1929 mA the result has become9.47 / 0.1929 = 49.611 Kilo-Ohm. A resistor of 47K would come close enough. Ofcourse you could also use a 50K trimpot to adjust the value even more accurately. The 1K5 (1500 Ohm) resistor in series with the LED is to limit the current thru the LED below 20 mA.

The only thing left is to calculate the value of the series resistor which determines the switch-over from charge to float condition. This occurs when the voltage drop over the 0.1 ohm (wire-wound) resistor at the positive leg smaller is than over the 2.2 ohm resistor. This value is 2.2 x 4.9564 = 10.9 mV. The resistance is 0.1 ohm, to get a voltage drop over this resistor of 10.9 mV is the current 10.9 x 0.1 = 109 mA. The second this charge current becomes lower then 109 mA, the LM1458 triggers over to the float condition.
The adjustment with the 100-ohm trimpot determines the maximum charge current. The voltage on the walker of this trimpot varies between 10.9 mV - 506.54 mV. The current is this way made adjustable between 0.1A - 5A, but we should not go that far because the LM350K can not handle anything over 3Amp. If we chose a trimpot with a value of 50 ohm, then on the other hand the 3A can not be obtained. So, careful adjustment is the remedy. Take your time!

With this information it is a simple task to calculate the dissipation values of the resistors. In other words, the product of the resistance multiplied with the current in square (I2xR).

The only resistor which gets it difficult is the 0.1 ohm, but then again, not by much 3 x 3 x 0.1 = 0.9 Watt.
Rest us to calculate the power. For that we have add a couple of voltages. We have the input voltage of 14.1, the voltage drop over the resistor, 0.1 x 3 = 0.33 volt, and 3 volt minimum over the LM1458 for proper function, total 17.43 volt. The transformer provides 18V (effective). With ideal rectifying this should total 18 x 1.41 = 25.38 volt. There are however losses via the diodes and bridge rectifier so there is about 23.88 volt remaining. Not much tolerance to play with, on the other hand, too much causes energy loss in the form of heat anyway.
The voltage drop over the buffer capacitor may not be lower than 17.43 volt, meaning, the ripple voltage may reach about 23.88 - 17.43 = 6.45 volt. By double-fase rectifying is the ripple voltage equal to I/(2xfxC) whereby I is the discharge current, f is the supply frequentie and C is capacity of the buffer capacitor in Farad. Exchanging places this would give C = 3/(2x50x6.45) = 0.004651 Farad, or 4651 uF. A standard value of 4700 uF with a minimum voltage value of about 35-40 Volt. The other capacitor is not very critical and is only there to kill small voltage spikes which could influence the operation of this charger otherwise.

The bridge rectifier gets a good workout also and it is therefore recommended to chose NOT a too light a unit. A 5A rectifier is often too small, better to take a 8 or 10A type. These are readily available everywhere.

Last but not least, the transformer. The buffer capacitor has approximately 25 volt accros. The current is 3A. This calculates to a power of 25 x 3 = 75 watt. This transformer has its own problems with powerloss (naturely occuring) and so a unit of about 80 watt is acceptable.
Never attempt to charge a 6 volt battery with a 12 volt charger; you are asking for trouble. Good luck all!

Foam Cutting Power Supply

2009-06-05T22:07:00.000-07:00

After seeing other modelers building their model wings from plastic foam, I decided that I wanted to do the same. Building your wings from foam covered with 1/16 in. balsa can produce a strong and light wing that could be difficult to duplicate with the standard balsa rib construction, especially if the wing had a duel tapered, symmetrical airfoil. The standard way to cut foam is with the Hot Wire technique, using steel or nichrome wire through which an electrical current flows to heat the wire.

However, the methods that many use to get the wire hot leaves something to be desired. The most common method I saw used was to connect a 12volt battery charger to 4 or 5 feet of nichrome wire which was tied to some kind of a bow. Using the variable charging rate, you could control (to a limited degree) the temperature of the wire and thus the speed of the cut. But if you cannot accurately control the heat, you'll get many poor cuts. Some have connected a series of light bulbs in line with the wall service of 115 volts AC.

It works, but WOW, is it ever dangerous! Terrible shock hazard! I've even seen some connect the nichrome wire across a 12 volt car battery, also very dangerous. Over the years there have been several schematics listed in the model magazine for building a hot wire foam cutter power supply. All of them worked, I'm sure. Some were very simple, but left little heat control, and others were complex and expensive. Heat control is the secret for making good foam cuts. Also a good transformer is important for removing the electrical shock hazard that threatens the modeler in his shop. A good current limiting feature also makes the device safe from high current burns, which some auto mechanics have suffered when working with large 12 volt batteries.

The following circuit is a simplification of several older designs. This design uses readily available parts, is easy to build, has total temperature control for both a long bow (48") and a short bow (24"), and has served me well for the last 15 years. Many of the planes that I fly are my own design and I build most of them with foam wings, foam turtle decks, foam stabs, etc, usually with dual tapered, symmetrical designs. The short bow is valuable for sculpting foam pieces into various shapes, as it can be held in one hand and the foam sample in the other.

The first step in building one of these foam cutters is to take the Bill of Materials to your local Radio Shack and search for the parts. I picked this source because of shopping convenience and the total cost is a little above $30. Also get a small copper clad circuit board (CB), about 3 by 4 inches or larger in size. If you chose not to make the circuit board, you can solder the parts together using electrical stand-offs. The first order of business is to mount the switches, the potentiometer, the Red and Black electrical posts (#274-662), and the red indicator light on the front panel of the component box according to the picture and illustration. Next mount the transformer (#273-1512), fuse holder (#270-364), and electrical cord (#278-1255) in the box as shown in photo. Put some rubber feet on the bottom of the box (also from Radio Shack) so that it won't scratch your wife's end table when you take it to show her what a great craftsman you are.

The circuit is a simple AC Triac voltage control circuit similar to the ones used to control house lamps. The transformer provides the electrical isolation the makes this item safe to operate. The voltage at the bow will tingle a little, but will not harm the operator. The OFF/ON switch is a simple s.p.s.t. switch (#275-651). The "Long/Short" Bow selector switch is the same part number. Across the primary side of the transformer is mounted an indicator "ON" lamp (#272-712) which will light up when the unit is turned on. The temperature control is through the 5k ohm potentiometer R2 (#271-1714). The Triac gate current is controlled by R3, a 470 ohm, 1 watt resistor. This resistor is not part of radio Shack's inventory, therefore it may be required to solder two 1k ohm, 10 watt resistors in parallel. The capacitor, C1, is a 0.22microF disk (#272-1070). The 5 ohm, 20 watt resistor R1 is made of two 10 ohm, 10 watt resistors in parallel. They are large ceramic resistors mounted side by side. These resistors drop the voltage when the short 24" bow is being used. These resistors will get hot, don't touch!

Enclosed in this article is a actual size drawing of the circuit board (CB, 2.5" x 4"). Cut out this drawing and use it as a template, and paste it on the side opposite of the copper on the CB (circuit board) with some rubber cement. Next use a center punch to mark the center of each hole. Then drill the holes with the CB held tightly to some wood backing, making the four corner holes a 1/8" in dia and all the rest about 1/16" in dia. These smaller holes will be where you solder the components and wires. The larger holes are for the mounting bolts to hold the CB to the case. Cut the CB to the exact size as shown on the template (2.5" x 4"). Then, print out the copper side drawing and paste it on the copper side of the board. Use a sharp X-acto knife to remove thin strips of copper as shown. This will isolate the copper soldering pads from one another. Remove the paper. Insert the components in the CB on the side opposite the copper. Where the component leads stick out on the copper side, solder the component leads to the board being careful not to allow solder to bridge the cut lines in the copper. Cut off any excessive lead after soldering it. Bolt the Heat Sink on to the Triac with the fins pointing out. The Triac should be mounted in a vertical position, perpendicular to the CB. Next mount the CB in the box with 6-32 x 1" bolts and stand-offs. Finish soldering the connecting wires to the board before tightening the bolts. Drill 3 or 4 vent holes (1/4" dia) in the top of the box in the area above the Triac heat sink.

Plug the nichrome wire bow leads into the dual plug speaker connectors. It is best to trim the wire insulation on the wires back about 1/2 in. then tin the wire ends. After the wires are plugged in, turn the unit on and with the temperature control at half point and the heat switch set up to long. The wire should get hot to the touch almost immediately. If it doesn't, then examine the construction on the circuit board and wiring, and fix any errors found. After the unit is finished, bolt the top on and your and you're done.

When you use the foam cutter, be sure that the Bow switch is in the correct position. The switch must be in the "Short" position (down) if the 24" bow is used. Otherwise you may blow the fuse. Leave the unit in the "Long" bow position at all times unless you are using the short bow and you should have no problems. Before you turn the Foam Cutter on, turn the temperature (TEMP) control fully counter-clockwise, to minimum temperature. Turn the unit on with a bow plugged in and increase the temperature by turning the TEMP knob clockwise. The temperature of the wire increases almost immediately. With a piece of foam, test for the foam cutting temperature. Reduce the TEMP control until the cut is smooth with little foam evaporation around the wire. Remember, the smoothest cuts are made slowly. Spend some time practicing until your cuts are smooth. You will never go back to balsa ribs!

Lead Acid Battery Charger

2009-06-05T21:03:00.000-07:00

Parts List:

C1 = 100uF/63V
C2 = 10uF/63V
D1 = 1N5401/NTE5801
D2 = LED (Red, 5mm)
Q1 = NTE374/BD140
Q2 = NTE123AP/BC547
R1 = 120 Ohm
R2 = 82 Ohm
R3 = 10K
R4 = 33K
R5 = 22K
P1 = 2K2
U1 = LM350 (On large coolrib!)

How it works:

Except for use as a normal Batter Charger, this circuit is perfect to 'constant-charge' a 12-Volt Lead-Acid Battery, like the one in your flight box, and keep it in optimum charged condition. This circuit is not recommended for GEL-TYPE batteries since it draws to much current.

The above circuit is a precision voltage source, and contains a temperature sensor with a negative temperature coëficient. Meaning, whenever the surrounding or battery temperature increases the voltage will automatically decrease. Temperature coëficient for this circuit is -8mV per °Celcius. A normal transistor (Q1) is used as a temperature sensor.

This Battery Charger is centered around the LM350 integrated, 3-amp, adjustable stabilizer IC. Output voltage can be adjusted with P1 between 13.5 and 14.5 volt. T2 was added to prevent battery discharge via R1 if no power present. P1 can adjust the output voltage between 13.5 and 14.5 volts. R4's value can be adjusted to accommodate a bit larger or smaller window. D1 is a large power-diode, 100V PRV @ 3 amp. Bigger is best but I don't recommend going smaller.

The LM350's 'adjust' pin will try to keep the voltage drop between its pin and the output pin at a constant value of 1.25V. So there is a constant current flow through R1. Q1 act here as a temperature sensor with the help of components P1/R3/R4 who more or less control the base of Q1. Since the emitter/base connection of Q1, just like any other semiconductor, contains a temperature coëficient of -2mV/°C, the output voltage will also show a negative temperature coëficient. That one is only a factor of 4 larger, because of the variation of the emitter/basis of Q1 multiplied by the division factor of P1/R3/R4. Which results in approximately -8mV/°C. To prevent that sensor Q1 is warmed up by its own current draw, I recommend adding a cooling rib of sorts.
(If you wish to compensate for the battery-temperature itself, then Q1 should be mounted as close on the battery as possible) The red led (D2) indicates the presence of input power.

Depending on what type of transistor you use for Q1, the pads on the circuit board may not fit exactly (in case of the BD140).

Caution: Adjust the voltage of capacitor C1 according to the input voltage. Example, if your input voltage will be 24 volt, your C1 should be able to carry at least 50V.

IC Controlled Emergency Light with Charger

2009-06-05T19:39:00.000-07:00

The circuit shown here is that of the IC controlled emergency light. Its main features are: automatic switching-on of the light on mains failure and battery charger with overcharge protection. When mains is absent, relay RL2 is in deenergised state, feeding battery supply to inverter section via its N/C contacts and switch S1. The inverter section comprises IC2 (NE555) which is used in stable mode to produce sharp pulses at the rate of 50 Hz for driving the MOSFETs. The output of IC3 is fed to gate of MOSFET (T4) directly while it is applied to MOSFET (T3) gate after inversion by transistor T2. Thus the power amplifier built around MOSFETs T3 and T4 functions in push-pull mode.

The output across secondary of transformer X2 can easily drive a 230-volt, 20-watt fluorescent tube. In case light is not required to be on during mains failure, simply flip switch S1 to off position.

Battery overcharge preventer circuit is built around IC1 (LM308). Its non inverting pin is held at a reference voltage of approximately 6.9 volts which is obtained using diode D5 (1N4148) and 6.2-volt zener D6. The inverting pin of IC1 is connected to the positive terminal of battery. Thus when mains supply is present, IC1 comparator output is high, unless battery voltage exceeds 6.9 volts. So transistor T1 is normally forward biased, which energises relay RL1. In this state the battery remains on charge via N/O contacts of relay RL1 and current limiting resistor R2. When battery voltage exceeds 6.9 volts (overcharged condition), IC1 output goes low and relay RL1 gets deenergised, and thus stops further charging of battery. MOSFETs T3 and T4 may be mounted on suitable heat sinks.

Precision Metronome & Pitch generator

2009-06-03T09:44:00.000-07:00

Precision Frequency generator 1 to 999 Hz
Precision Metronome 1 to 999 beats per minute

Circuit Diagram:

Parts:
R1__________1M 1/4W Resistor
R2_________22K 1/4W Resistor
R3__________6K8 1/4W Resistor
R4__________4K7 1/4W Resistor
R5_________47K 1/4W Resistor
R6________100K 1/4W Resistor
R7_________39K 1/4W Resistor
R8_________12K 1/4W Resistor
C1_________47pF 63V Ceramic Capacitor
C2_______2-22pF 63V Ceramic Trimmer
C3________470pF 63V Ceramic Capacitor
C4_________10pF 63V Ceramic Capacitor
C5________100nF 63V Polyester Capacitor
C6________220nF 63V Polyester Capacitor
C7_________22µF 25V Electrolytic Capacitor
D1-D15___1N4148 75V 150mA Diodes
IC1________4060 14 stage ripple counter and oscillator IC
IC2________4082 Dual 4 input AND gate IC
IC3________4520 Dual binary up-counter IC
IC4________4518 Dual BCD up-counter IC
IC5________4046 Micropower Phase-locked Loop IC
IC6________4040 12 stage ripple counter IC
Q1________BC337 45V 800mA NPN Transistor
XTAL______2.4576 MHz Miniature quartz crystal
SW1__________BCD Miniature Thumbwheel Switch (units)
SW2__________BCD Miniature Thumbwheel Switch (tens)
SW3__________BCD Miniature Thumbwheel Switch (hundreds)
SW4_________SPST Slider Switch (On-off)
SW5_________SPDT Slider Switch (Metronome-Pitch)
SPKR_______8 Ohm, 50 mm. Loudspeaker
B1_________9V PP3 Battery
Clip for 9V PP3 Battery

Circuit operation:

CMos IC1 and IC2B quad AND gate form a 2.4576 MHz crystal oscillator plus a 2400 times divider. IC3A provides further division by 16, delivering a 64 Hz stable frequency square wave. This frequency is multiplied (by means of Phase Locked Loop IC5, double decade divider IC4 and IC3B 4 bit binary divider) by the number set by three miniature BCD thumbwheel switches SW1, SW2 and SW3: units, tens and hundreds respectively.

Connecting, by means of SW5, Q1 base to pin 2 of IC6, we obtain (after a 64 times division) the same frequency set by thumbwheel switches with quartz precision, and no need for a scale indicator.

Volume regulation of the pitch generator is obtained trimming resistor R5. In the same way, with SW5 set to metronome, the small speaker reproduces the frequency set by thumbwheel switches but divided by 3840, thus obtaining beats per minute ratio.

Fuzz-box

2009-06-03T09:36:00.000-07:00

All-FET design Valve-like distortion behavior

Parts:

P1______________10K Log. Potentiometer
R1_______________1M 1/4W Resistor
R2_______________3K3 1/4W Resistor
R3_______________2K2 1/4W Resistor
R4_______________5K 1/2W Trimmer (Cermet)
R5_____________100K 1/4W Resistor
C1,C4__________100nF 63V Polyester Capacitors
C2_____________100pF 63V Ceramic Capacitor
C3,C5___________22µF 25V Electrolytic Capacitors
Q1,Q2,Q3______2N3819 General-purpose N-Channel FETs
J1,J2__________6.3mm Mono Jack sockets
SW1_____________DPDT Toggle - Slider or Pedal Switch
SW2_____________SPST Toggle or Slider Switch
B1________________9V PP3 Battery
Clip for PP3 Battery

Comments:

This circuit was designed to obtain a valve-like distorted sound from an electric guitar or other musical instrument.

For this purpose a very high gain, three-FET amplifier circuit, was used. The output square wave shows marked rounded corners, typical of valve-circuits when driven into saturation.

Therefore, the distorted sound obtained from such a device has a peculiar tone, much loved by most leading guitarists.

Technical data:

Input sensitivity: 30mV RMS.
Output square wave: 6V peak-to-peak max.
Total current drawing: about 1mA.

Circuit set-up using oscilloscope and sine wave generator:
Connect a 1KHz sine wave generator to J1 and the oscilloscope to J2.
Adjust R4 until the output square wave shows equal mark-space ratio.

"By ear"

circuit set-up:

Connect a musical instrument to J1 and an amplifier to J2.
Carefully adjust R4 in order to obtain as maximum output sound intensity as possible.

UltraSonic Radar

2009-05-26T19:15:00.000-07:00

This is a very interesting project with many practical applications in security and alarm systems for homes, shops and cars. It consists of a set of ultrasonic receiver and transmitter which operate at the same frequency. When something moves in the area covered by the circuit the circuits fine balance is disturbed and the alarm is triggered. The circuit is very sensitive and can be adjusted to reset itself automatically or to stay triggered till it is reset manually after an alarm.

How it Works

As it has already been stated the circuit consists of an ultrasonic transmitter and a receiver both of which work at the same frequency. They use ultrasonic piezoelectric transducers as output and input devices respectively and their frequency of operation is determined by the particular devices in use.

The transmitter is built around two NAND gates of the four found in IC3 which are used here wired as inverters and in the particular circuit they form a multivibrator the output of which drives the transducer. The trimmer P2 adjusts the output frequency of the transmitter and for greater efficiency it should be made the same as the frequency of resonance of the transducers in use. The receiver similarly uses a transducer to receive the signals that are reflected back to it the output of which is amplified by the transistor TR3, and IC1 which is a 741 op-amp. The output of IC1 is taken to the non inverting input of IC2 the amplification factor of which is adjusted by means of P1. The circuit is adjusted in such a way as to stay in balance as long the same as the output frequency of the transmitter. If there is some movement in the area covered by the ultrasonic emission the signal

that is reflected back to the receiver becomes distorted and the circuit is thrown out of balance. The output of IC2 changes abruptly and the Schmitt trigger circuit which is built around the remaining two gates in IC3 is triggered. This drives the output transistors TR1,2 which in turn give a signal to the alarm system or if there is a relay connected to the circuit, in series with the collector of TR1, it becomes activated. The circuit works from 9-12 VDC and can be used with batteries or a power supply.

Construction

First of all let us consider a few basics in building electronic circuits on a printed circuit board. The board is made of a thin insulating material clad with a thin layer of conductive copper that is shaped in such a way as to form the necessary conductors between the various components of the circuit. The use of a properly designed printed circuit board is very desirable as it speeds construction up considerably and reduces the possibility of making errors. Smart Kit boards also come pre-drilled and with the outline of the components and their identification printed on the component side to make construction easier. To protect the board during storage from oxidation and assure it gets to you in perfect condition the copper is tinned during manufacturing and covered with a special varnish that protects it from getting oxidised and also makes soldering easier. Soldering the components to the board is the only way to build your circuit and from the way you do it depends greatly your success or failure. This work is not very difficult and if you stick to a few rules you should have no problems. The soldering iron that you use must be light and its power should not exceed the 25 Watts. The tip should be fine and must be kept clean at all times. For this purpose come very handy specially made sponges that are kept wet and from time to time you can wipe the hot tip on them to remove all the residues that tend to accumulate on it. DO NOT file or sandpaper a dirty or worn out tip. If the tip cannot be cleaned, replace it. There are many different types of solder in the market and you should choose a good quality one that contains the necessary flux in its core, to assure a perfect joint every time. DO NOT use soldering flux apart from that which is already included in your solder. Too much flux can cause many problems and is one of the main causes of circuit malfunction. If nevertheless you have to use extra flux, as it is the case when you have to tin copper wires, clean it very thoroughly after you finish your work. In order to solder a component correctly you should do the following:

Clean the component leads with a small piece of emery paper.
Bend them at the correct distance from the component�s body and insert the component in its place on the board.
You may find sometimes a component with heavier gauge leads than usual, that are too thick to enter in the holes of the p.c. board.
In this case use a mini drill to enlarge the holes slightly. Do not make the holes too large as this is going to make soldering difficult afterwards.
Take the hot iron and place its tip on the component lead while holding the end of the solder wire at the point where the lead emerges from the board. The iron tip must touch the lead slightly above the p.c. board.
When the solder starts to melt and flow wait till it covers evenly the area around the hole and the flux boils and gets out from underneath the solder. The whole operation should not take more than 5 seconds. Remove the iron and allow the solder to cool naturally without blowing on it or moving the component. If everything was done properly the surface of the joint must have a bright metallic finish and its edges should be smoothly ended on the component lead and the board track. If the solder looks dull, cracked,or has the shape of a blob then you have made a dry joint and you should remove the solder (with a pump, or a solder wick) and redo it.
Take care not to overheat the tracks as it is very easy to lift them from the board and break them.
When you are soldering a sensitive component it is good practice to hold the lead from the component side of the board with a pair of long-nose pliers to divert any heat that could possibly damage the component.
Make sure that you do not use more solder than it is necessary as you are running the risk of short-circuiting adjacent tracks on the board, especially if they are very close together.
When you finish your work cut off the excess of the component leads and clean the board thoroughly with a suitable solvent to remove all flux residues that may still remain on it.
There are quite a few components in the circuit and you should be careful to avoid mistakes that will be difficult to trace and repair afterwards. Solder first the pins and the IC sockets and then following if that is possible the parts list the resistors the trimmers and the capacitors paying particular attention to the correct orientation of the electrolytic.

Solder then the transistors and the diodes taking care not to overheat them during soldering. The transducers should be positioned in such a way as they do not affect each other directly because this will reduce the efficiency of the circuit. When you finish soldering, check your work to make sure that you have done everything properly, and then insert the IC�s in their sockets paying attention to their correct orientation and handling IC3 with great care as it is of the CMOS type and can be damaged quite easily by static discharges. Do not take it out of its aluminium foil wrapper till it is time to insert it in its socket, ground the board and your body to discharge static electricity and then insert the IC carefully in its socket. In the kit you will find a LED and a resistor of 560 � which will help you to make the necessary adjustments to the circuit. Connect the resistor in series with the LED and then connect them between point 9 of the circuit and the positive supply rail (point 1).

Connect the power supply across points 1 (+) and 2 (-) of the p.c. board and put P1 at roughly its middle position. Turn then P2 slowly till the LED lights when you move your fingers slightly in front of the transducers. If you have a frequency counter then you can make a much more accurate adjustment of the circuit. Connect the frequency counter across the transducer and adjust P2 till the frequency of the oscillator is exactly the same as the resonant frequency of the transducer. Adjust then P1 for maximum sensitivity. Connecting together pins 7 & 8 on the p.c. board will make the circuit to stay triggered till it is manually reset after an alarm. This can be very useful if you want to know that there was an attempt to enter in the place which are protected by the radar.

Componets:

R1 = 180 KOhm
R2 = 12 KOhm
R3, 8 = 47 KOhm
R4 = 3,9 KOhm
R5, 6, 16 = 10 KOhm
R7, 10, 12, 14, 17 = 100 KΩ
R9, 11 = 1 MOhm
R13, 15 = 3,3 KOhm
C1, 6 = 10uF/16V
C2 = 47uF/16V
C3 = 4,7 pF
C4, 7 = 1 nF
C5 = 10nF
C8, 11 = 4,7 uF/16
C9 = 22uF/16V
C10 = 100 nF
C12 = 2,2 uF/16V
C13 = 3,3nF
C14 = 47nF
TR1, 2, 3 = BC547 , BC548
P1 = 10 KOhm trimmer
P2 = 47 KOhm trimmer
IC1, 2 = 741 OP-AMP
IC3 = 4093 C-MOS
R = TRANSDUCER 40KHz
T = TRANSDUCER 40KHz
D1, 2, 3, 4 = 1N4148

Port-Powered Temperature Meter

2009-05-12T02:27:00.000-07:00

This is a four-channel temperature measurmet adapter that works without external power supply. It will suitable for measureing temperature and logging its data with a PC. The circuit diagram is very simple and no adjustment is required, everybody will able to build it with ease :-)

Specs. Micro-controller ATtiny15L (Atmel)
Number of channels Four channels
Measurement Range -40°C to +105°C (0.1°C/step)
or raw A-D value
Measurement Error ±0.5°C (at room temperature)
Sensor 103AT (Semitec)
Power Supply Supplyed from COM port (typ. 5mA)
Cost Approx. 1200JPY (All parts)

HARDWARE

Micro-controller

I chose an Atmel ATtiny15L for this project. It is the only device that has a built-in 10bit A-D converter in the 8 pin AVRs. The A-D converter has a bandgap reference and differencial amplifire as its front-end. The AVR core is clocked by only internal RC oscillator (calibrated to 1.6MHz), any other clock souce cannot be used. Also 25.6MHz clock source that 16x multiplied from core clock is available for timer/counter. This means that a fast PWM output can be generated. Therefore the ATtiny15L has good analog I/O capabiltity.

In this project, the A-D converter is used as four channels, single-ended, no gain and VREF from Vcc configuration. However RSTDISBL fuse must be programmed in order to use pin #1 as one of the analog inputs, an AVR programmer that can program in HVS mode is required.

Power Supply

The devices that works on the COM port without external power supply, such as serial mouse, are powered from the COM port. When an application program opens COM port, ER and RS signals will go high. The high level voltage is from 6V to 12V at most PCs, and it can supply 5mA at least. This is sufficient for low power micro-controllers.

Sensors

Four 103AT precision thermisters are used as temperature sensor. Its variation is very small, its temperature - resistance error at room temperature is ±0.3°C. The error of series resisters should be within ±0.5% to enable calibration-free design.

FIRMWARE

The program only respond the values of each channel to the PC by trigger command. The temperature - resistance curve of the thermister is not linear so that the raw A-D value is linearlized and converted to temperature value in software process. When replace the thermister with any oters, the linearlization table in the source code must be re-built. The raw A-D value can also be read, it will be used as voltage meter.

The trigger command is one "T" or "R" character, returened results are the temperarute for "T" command, raw A-D value for "R" command. Each value is separated by a comma and terminated by a .

Free Firmware

Digital Capacitance Meter

2009-05-11T23:22:00.000-07:00

This is a simple capacitance meter which can measure capacitance value easy. There are some measurement methods for capacitance, at one time the capacitance was measured with a impedance bridge or a dip meter. Recently typical capacitance meters can measure capacitance and some additional characteristics from current vector by applying AC voltage to the Cx. Some simple capacitance meter use integration method that measureing transient response of the R-C network. There are some construction kits based on this method.

This project uses the integration method. There is an advantage that the resulut can be got as a digital data directly because it bases measurement of time, accurate analog circuit is not required and its calibration can be done easy by using a micro controller. Therefor the integration method is suitable for hand built capacitance meter with high realizability.

Hardware

To measure a charge time, only a voltage comparator, a counter and some glue logics are needed. However, a microcntroller (AT90S2313) is used for this project to realize the system easy. In point of fact, I had thought that analog comparator in the AVR is not useful. But I found that the compare output can also be used as a captureing trigger of TC1. This is a nice feature for that use :-)
The integration circuit can be simplified like shown in the circuit diagram. The threshold voltage is generated by divider registers. It seems not stable to valiation of supply voltage however the charge time is not affected by the supply voltage. You will able to find that voltage terms can be erased when apply formure 2, VC1/E term is dtermined by only divide ratio. This advantage is the essence found in the NE555 timer IC. Ofcourse the supply voltage must be steady during integration.

According to the foundation, measure integration time with only one threshold voltage will do. However input voltage of near ground level is little difficult to use due to following reasons.

Voltage not drop to 0 volt. Capacitor voltage will not be discharged to zero volt. It require a time to discharge capacitor to sufficientaly low voltage for measuring operation. It will expand measureing interval. Saturation voltage at discharge switch is also increase this effect.
There is a time beween start to charge and then start timer. It will cause a measurement error. This can be ignored on the AVR because it requires only one clock cycle for that sequence. Any other microcontroller may rquire to consider this problem.
Leakage current on analog input. Accrding to AVR data sheet, the leakage current on analog input is increased near zero volt. This will cause a measurement error.

To avoid to use near zero volt, two threshold voltages VC1(0.17 Vcc) and VC2(0.5 Vcc) are used and measure t2-t1(0.5RC). This can avoid avobe problems and comparator delay/offset will also be canceled. As for the leakage currnet, circuit board should be kept clean to minimize surface leak.

The supply voltage is generated with a DC-DC converter powered from a 1.5V AA cell. The swiching power supply is not suitable for measurement circuit but it seems not affected by ripple voltage because two ripple filters are applied. I recommend to use a 9V 6LR61 battery and a 78L05 instead, and do not omit BOD or you will be afflicted with EEPROM data collaption.
Calibration

When power is on first time, full segment, "E4" and ten several pF will be displayed. This value means stray capacitance on the circuit. The stray capacitance can be canceled by SW1. Two reference capacitors of 1nF and 100nF are needed to calibrate the capacitance meter. If you could not obtain the reference capacitors, accurate capacitors within ±1% can be used insted. This capacitance meter does not have any trimmer pot, it performs the calibration by reading the reference capacitor and saving gain adjustment value in full automatic operation.

To calibrate low range: First, adjust zero with SW1. Next, tie pin #1 and #3 of connector P1, set a 1nF reference capacitor and push SW1.

To calibrate high range: Tie pin #4 and #6 of connector P1, set a 100nF reference capacitor and push SW1.

"E4" at power on means calibration value in the EEPROM has been broken. It will never be displayed if once calibration is performed. As for zero adjustment, it is not saved into the EEPROM, it will require each time power-on or any jig is attached.

Free Firmware

Radio Spectrum Monitor

2009-05-10T19:10:00.000-07:00

This is an experimental work to monitor a spectrum pattern in radio band, and is a continuous project from Audio Spectrum Monitor. To analyze the spectrum of an input signal, I chose an Atmel AVR micro controller that used in the Audio Spectrum Monitor to process FFT. When think it easy, it can be thought that sample an input RF signal directly and analyze it will do. However, you will able to recognize that there are some technical difficulties from following reasons.

The acquisition unit must have a sufficient speed and accuracy that covering over the radio frequency range. As for AM radio band like this project, fast 12 bit ADC and specific controller will able to cover this range. However there is no proper ADC for UHF band.
Number of samples to meet required frequency resolution, fSAMP/fFUND samples, becomes too large. When monitor an AM radio band around 1 MHz in frequency resolution of 500 Hz, over 4000 samples will be required at least. And when monitor a VHF band in same frequency resolution, how many samples will be required...

To solve this difficulties, there is a generic method called Frequency Conversion. In brief, down-convert the RF signal to lower frequency with a mixer (multiplyer) before sampling stage. When process the signals as complex signal for the frequency conversion, it can handle negative frequency, and the center frequency can be moved to zero hertz without interference by image signals. This means that the sampling frequency higher than span frequency range will do. When monitor a 100 MHz band in span of ±1 MHz, it will be converted down to 0±1 MHz and sampled it in only 2 Msps. You may able to understand easy when explain it as `cut and paste' on the frequency domain. Actually, this kind of radio spectrum monitors are being supplied from some radio equipment vendors.

Hardware

In this project, an intermediate frequency signal from mixer output of an AM radio (fC=455kHz) is used as an input signal. It is converted again down to zero hertz in complex signal, so that the signal path, mixer and local oscillator, must be composed for complex signal processing. The complex signal is expressed in two wire IQ signal, the real part corresponds to I signal and the imaginary part corresponds to Q signal. The arithmetic circuits for IQ signal are realized in method of complex arithmetic. For example, a mixing circuit for IQ signals requires four multipliers and two adders from the formula:

(a1+jb1)(a2+jb2) = (a1a2-b1b2)+j(a1b2+a2b1)

Therefore the analog process of IQ signal requires large scale circuit compared to real signal process, so that the IQ signal is usually digitized after minimal analog process and following process are done in digital processor. In case of one input is real, only two multiplyer will do. By the way, when view a real signal as a complex signal, its spectrum pattern of positive part and negative part is line symmetrical. Real signal can be saied that the imaginary part is always zero. The case of complex signal becomes such state is: there are conjugate complex numbers, changed sign of the imaginary part (changed sign of frequency), for each complex frequency components. Therefore the real signal has symmetrical spectrum around the origin.

Software

The firmware samples IQ signal, analyze it in FFT algorithm and draw spectrum pattern into LCD module. These processes are done in refresh rate of approximately 60 times per second. IQ signal is sampled 128 points in samplig rate of 64 ksps at a time, span frequency of ±32 kHz around local frequency (455 kHz) can be monitored. Now horizontal scale (frequency) is labeled on the LCD but when measure local frequency of the radio receiver, the tuned frequency will able to be displayed under the spectrum bars - I became aware this idea during writing this document and implemented it in a hurry:-) When a local input from receiver is valid, frequency scale is appered in the LCD and tuned frequency is refrected.

The fast FFT routine is copied from audio spectrm monitor as is. Basically, FFT algorithm is in complex input/output, it can be used for complex signal with no modification unless it is optimized for real input. When input is a real signal, only half of result is valid because it is symmetrical around the origin. Ofcourse each domain of output is valid when input is a complex signal.

When apply a power, a menu window will apper and can be set working condition with a joystick. Each item can be selected with up/down action, changed with right action and enter running mode with push action. In running mode, up action hold/resume display, down action resets peak hold (if selected), left action redo automatic null (offset cancellation of ADC input) and push action returns to menu. Window function can also be selected, you will able to recognize difference between each window. Wave form mode monitors raw IQ signal but it will not useless.

Free Firmware

Adjustment

Adjust local frequency to 455 kHz with TC1.
Apply 455+10 kHz, 100mVP-P sin wave to the input port and set I and Q signal as same amplitude and quadrature phase at ADC input with VR1 and VR2. And confirm that DC levels are Vcc/2 (a little offset is negligible) and no distortion is recognized.
Enter running mode. When a spectrum bar is 20 bins right from center, it is working successfully. If an image is appering at opposite position, adjust VR1 and VR2 to eliminate it carefully. A peak appearing center at start is due to left DC offset of IQ signal, it will be nulled automatically.

An AM super heterodyne radio receiver is used as a signal source. It may be modified to export mixer output and local oscillator output (this can be ommited). When it is a transister radio, large local frequency component (tuned freqency + IF) will pass through the mixer due to the mixer will be cheap emitter injection type. This affects dynamic range of DBM input, so that unnecessary signals above IF must be filtered out with a LPF. When IF frequency of radio receiver is 450 kHz, please read "455" in this page to "450".

GPS Data Logger

2009-05-10T18:47:00.000-07:00

I have got a GPS module last year and I built a GPS data logger that records position data from the GPS module. The position data is output in NMEA-0183 format and store its sentence into any storage device. The position data can be processed with existing GPS utilities for interesting applications, such as Tracing the route on Google Maps (HTML source text).

Hardware

This is the inside of the built data logger and the circuit diagram. To store the tracking log, MMC/SDC is used for the recording media. The MMC/SDC is the most suitable purpose to collect the logged data to the PC. The GPS data logger is mainly used on automobile, so that its power supply circuit should pass the load dump surge immunity test. The operating power is got from only ACC line for good usability, and power switch is ommited. To detect brown-out and power-off, input supply voltage is monitored with the microcntroller. The controller part works at 3.0 volts and the GPS module works at 5.0 votls are tied via a level converters each other.

Software

When a positioning is established and a valid RMC sentence is detected, logging operation is started with a log file named in current UTC time YYMMDD.log. When same name is already existing, it starts to store from end of the file.

Because the operating power is got from ACC line, power-off will occure asynchronously regardless of the operating state. When power-off is occured, open file in write mode must be closed quickly during operating power is retaining in the capacitor, or the log file will be lost. In this project, when supply voltage is continuously below 8.0 volts for 10 milliseconds, it reecognizes that power-off and close the log file.

Free Firmware

Graphic MP3 Player

2009-05-07T23:03:00.000-07:00

The MP3 players consist of only semiconductor parts and no complex mechanics. This is a great feature for electronics handicrafts because it can be built easy with the same performance as commercial products except for the appearance. After a long blank in MP3 project, I built a new one again as the second MP3 project.

The first MP3 project was Pocket sized MP3 Player built with an MP3 chipset that obtained by chance. I have completed some projects of MP3 application for my business but not for hobby because I have no practice of listening music at the outdoors. Why did I built it in pocket size? Because that was only an impulse and nothing else. BTW, the MP3 player is being used in my car as a car MP3 player :-)

Now, there are many easy-to-use MP3 decoders that integrates DSP, DAC and amplifier on a chip. As the result, the MP3 player becomes to a popular project for electronics handiworks and everybody is enjoying to build it as their original project. One day I got to want to build an MP3 player by a reason (described below) and decided to start a new project. This is a regular project on the MP3 player after 8 years. I designed it as a desktop player because portable player is not useful for me.

Hardware

Below image shows the block diagram and the circuit diagram of built MP3 player. It has a feature that it has a large color LCD and a touch screen. Followings describe on each block.

Controller

A V850ES/JG2 (NEC Electronics) is used for system control. This is a 32-bit RISC microcontroller with 256KB flash and 24KB RAM. It was not that well known for electronics handiworks but somebody will be interesting in it because the V850 board was bundled as supplement of magazine in this year. For ordinary MP3 players, most 8-bit microcontroller is sufficient to build it. However this project requires a microcontroller with external memory interface because the controller must handle large amount of image data. Of course any popular microcontrollers, such as Renesas SH2 and H8, will able to be used as well. The reason why I chose the V850 is from its very low power consumption and the serial interface is easy-to-use better than Renesas's one. The V850ES/JG2 can run at 20 MHz but it is used at 14.7 MHz (4xPLL from 3.68 MHz xtal) to deliver a clock signal to MP3 decoder and LCDC.

Storage Media

SD Memory Card is the de facto standard of flash memory card. It can be attached to the microcontroller via a few signal lines. FAT format is used to store data files in it so that the project using the memory card must implement the FAT file system. Fortunately, there are various FAT libraries on the web as freeware so that everybody can use the memory card in their project with ease.

LCD Module

A 4″ color STN-LCD module in resolution of 320x240 is used for the display. Recently the price of color TFT-LCD are going falling and color STN-LCD will soon be shut out of the market. This project was started to to use this LCD module before it decays in the junk box.

A touch screen is attached on the LCD module, so that command buttons can be omitted. A CCFL is used for the back light and a CCFL inverter is required to drive the back light.

LCD Controller

The LCD module in this degree of resolution with built-in display buffer is not available and it must be refreshed by external circuit like CRT display system. Therefor it requires an additional LCD controller on the board to drive the graphic LCD module. The display buffer (RAM) is integrated in the LCDC or attached externally. In this project, an S1D13705 (EPSON) is used for LCD control. The S1D13705 has 80KB integrated display buffer and it can display in resolution of 320 by 240 with 8-bit color depth (256/4096 indexed color). It can be attached to the host controller via a 16-bit SRAM like interface.

The LCD module requires a 3.3V logic supply and an LCD bias supply (21-25V/3mA). Generally, the contrast of the STN-LCD in high drive duty ratio is affected by ambient temperature, so that a contrast dial is required to adjust the contrast. The contrast dial varies the LCD bias voltage. In this project, the contrast is adjusted automatically in software with a thermister put on the LCD module and a D-A converter.

MP3 Decoder

Recently VS10xx family (VLSI Solutions) is used for most home-built MP3 projects because it is easy to obtain and use. The VS10xx is designed for portable audio equipments and can drive a headphone directly. However it has only analog outputs and no digital (I2S) output, so that the VS10xx is not good when require an I2S output to attach an external DAC besides the analog performance is not good for Hi-Fi audio.

I chose STA013 (ST Microelectronics) for MP3 decoding. The STA013 have been released at the dawn of the MP3 format and widely used as a well known MP3 decoder chip. It has only digital (I2S) output, so that a proper audio DAC is required. This is an advantage on electronic handiworks because it can use various DAC chips and output audio data in SPDIF format with DAI encoder.

The STA013 has an integrated PLL oscillator to generate an audio timing clock (384fs) depends on DSP clock. If there is a jitter on the sampling clock, SNR of the analog output will be worse especially on Sigma-Delta DAC. Therefore the PLL power should be tightly filtered and pay attention to parts layout around the loop filter.

Analog Block
A PCM1748KE (BurrBrown) is used. The analog performance on the data sheet is not so bad but it is a little difficult to achieve expected performance because it is a Sigma-Delta DAC.

A post-filter is required at DAC output but only a slow roll-off one will do due to integrated 8x over sampling digital filter. In this circuit, the DAC output is filtered with a LPF+buffer and then output it as a line output and there is no speaker out. Therefore the MP3 player is used with any audio power amplifier.

The line output is tied to ADC input of the microcontroller. This is to get amplitude of the line output and display it as a level meter on the LCD. Generally an envelope detector is used for the audio level meter. In this project, to eliminate the envelope detector, the microcontroller samples waveform in sampling rate of 1kHz and detects peak-to-peak value from 20 samples. There will be dip points on the frequency response but there is no problem because this is for only a visual effect.

Building the MP3 Player

The circuit board must be embedded into a space in height of 10mm, so that the allowable height of the components on the circuit board is less than 6mm. Most of components used in this project ware surface mounted device. They are mounted on the proto-board directly and wired with UEW. This method requires soldering skill and practices but there is an advantage that it can achieve the density of double layered PCB or more. The FPC connector (0.5mm) easily creates solder bridge due to its terminal forms, so that solder the wire to the terminal via a stripe PCB instead of solder the wire directly.

The built circuit board is embedded with the LCD module into the case. The case is a clear acrylic case SK-16 (110x78x32mm) sold from Akizuki. Its depth was too long for this project, so that I cut down it to 25mm, paint black from inside and put an aluminum sheet for electromagnetic shield.

Free FirmWare

YUV(YCbCr) to RGB converter

2009-05-07T18:44:00.000-07:00

Recently, most digital video equipments, such as video recorder, DVD player and TV game, have component video output. The component video signal is like RGB video signal, but it cannot connect to RGB monitor directly. Thus I designed and built YUV(YCrCb) to RGB converter to use old TV monitor or video equipment which does not have component video input.

For old RGB monitor....no, my principal aim is to generate RGB signal from component signal from PlayStation 2 for scan converter because recent shipped PlayStation 2 fixes configuration of video output format to component video when used as DVD player :-( Converting the component signal into RGB signal can enjoy the game and DVD with high quality picture. However, the PlayStation 2 can be fixed to RGB output configuration with only a jumper wire. If you wish to get only RGB signal from the PlayStation 2, I recommend this way instead :-)

The theory of RGB to YUV conversion

The component video signals are generated by separating a luminance component and two chrominance components from RGB signals with following formulae:

Y = 0.299R + 0.587G + 0.114B Luminance component

R-Y = R - (0.299R + 0.587G + 0.114B)
= 0.701R - 0.587G - 0.114B Chrominance component (Red)

B-Y = B - (0.299R + 0.587G + 0.114B)
= -0.299R - 0.587G + 0.886B Chrominance component (Blue)

Note: These parameters are for SDTV(525/625), not for HDTV(750/1120).

G-Y crominaice component can also be generated. However, to restore the RGB signal, Y and two chrominance components will do. The G-Y component contains least chrominance in the three chrominance components so that this term is omitted to minimize conversion error.

When restore RGB signals from component signals, following formulae are applied.

R = Y + (R-Y)
G = Y - 0.51(R-Y) - 0.186(B-Y)
B = Y + (B-Y)

The contents of component video signal are these three signals, Y, R-Y and B-Y. You will able to understand that these are loss-less conversion. However, in order to reduce overall video signal band-width to be recorded or transmitted, the band-width of chrominance components are reduced and some compression processes are applied. Human eyes are sensitive to luminance but not sensitive to chrominance. The quality of chrominance components, such as band-width, SNR and digitizing resolution, can be reduced compared with luminance component.

The chrominance signal levels at the transfer line are normalized to luminance amplitude because it is easy to A-D, D-A conversion and transmittion processes. Following are attenuation ratio for the chrominance signals:

Cr = 0.713(R-Y)
Cb = 0.564(B-Y)

And sync pulse is added to luminance signal as -0.3V pulses. These are the component video signals that is appering at input/output point. Typical signal levels at the interface connector are from -0.3V to +0.7V for luminance signal, ±0.35V for chrominance signals.

Hardware

To convert component video signals into RGB video signals, the alithmetic circuit composed with some OPAMPs is used. However, only it is not sufficient to build complete set, some glue logics are also required.

Most video signal outputs except some high end equipments are AC coupled, DC restore circuit is needed to video input part. To remove sync pulses added to Y signal, blanking circuit is also required. These function should apply for completed process. The timing generator for the functions is realized with only a CPLD because to realize it with discrete components will be complex. Following images show the generated wave form of blanking pulse and clamping pulse. Top one is Y input, below two are /BLANK and /CLAMP. Left image is near horizontal blanking area, right image is near vertical blanking area.

Relationship of picture quality between component video and its source

Difference between component video signal and YC video signal is the transfer method of chrominance components.

At YC video signal, a color sub-carrier is modulated by two chrominance signals with quadrature balanced modulation, and transferred as a chrominance signal. Therefore, the band-width of chrominance signals are reduced to half of the sub-carrier frequency. At NTSC video format, I,Q signal which is shifted 33 degree from R-Y,B-Y axis is used instead of R-Y,B-Y signal, the band-width for I signal is 1.5 MHz, for Q signal is 0.5 MHz. However, most video decoders seem decode in R-Y,B-Y axis and both signals are limited to 0.5 MHz.

At component video siglal, two chrominance signals are transferred with two separated lines directly. This is full band-width transmittion. When video signal is from TV game whose source is RGB video buffer, difference between component signal and YC signal appers conspiculusly as some effects shown in following images. In this case, component signal is better than YC signal.

NTSC Test Signal Generator

2009-05-07T01:22:00.000-07:00

A video test pattern generator will be needed when calibrate or test video equipments. To characterize system performance and understand it quantitative, wave form monitor, vector monitor and related video measurement equipments will be required. Some generic patterns, shch as color bars, cross hatch, dots and full field, are also used for home video creation to calibrate video monitors without any measurement equipment.

This kind of test signal generators for personal use are often found as construction kits, however, most of the kits are designed on cheap analog method and the wave form is not accurate, therefore it cannot be used as a standard signal in fact. I designed and built a digital video test signal generator in order to generate accurate reference signals which can be used for calibration and measurment of video montors and home built video equipments.

Basic Design

Supported test patterns

The test signals used for electronics handiworks are required at least, plus some additional test signals are also implemented. However most test signals are used with wave form monitor, vector monitor, spectrum analyzer and others. An oscilloscope will able to be used instead of the wave form monitor in most case.

Color video signal format

This signal generator uses a three channel DAC to generate Composite video signal (CVBS) and S video signal (Y/C separated) at the same time, left one channel is not used in NTSC format. It is assigned for one of the color components of Y/CB/CR video format. This feature was not planned some years ago, but it has been added when started to draw the schematic this year, because NTSC television system might be obsoleted in the near future. The two different video format, component video and NTSC video, work in exclusive.

Specifications

Used LSIs DAC: MB40988 (Fujitsu)
MCU: ATmega161 (ATMEL)
PLD: XC95108 (Xilinx)
Output terminals CVBS: RCA pin jack
Y/C: 4pin Mini-DIN (S1)
Y/CB/CR: 14pin D type half pitch connector (D2)
Output impedance Video out: 75 ohm
ID out: 10 kohm
NTSC output
D1 output (480i) Timing: SMPTE 170M (RS-170A), Setup=0IRE (NTSC-Japan)
CVBS output: Y=714mV, Sync=-286mV, Burst=286mVp-p
Y/C output: Y=714mV, Sync=-286mV, Burst=286mVp-p
Y/CB/CR output: Y=700mV, Sync=-300mV, CB/CR=±350mV
D2 output (480p) Timing: SMPTE 293M
Y/CB/CR output: Y=700mV, Sync=-300mV, CB/CR=±350mV
Sampling 8bit x 3, 28.63636MHz
Power Supply AC100V 50/60Hz, <5w style="font-weight: bold;">

Hardware

Genarating a video signal

To generate any video signal, generic frame memory architecture will be used at most case. All samples of the video frame are stored into the frame memory and they are read in sequence and fed to video DAC. The difference between video signal generator and PC's video system is: which is stored into the frame memory, whole wave form including blanking/sync pattern or only pixel values in visible area.

The samplig frequency is 4 fsc (approx. 14.3MHz) at least. This is the typical value on the NTSC video system. However when use 4fsc for video measurements, an excellent video filter which has very sharp cut off, flat group delay and aperture effect compensation, is requied. This is very expensive and difficult to obtain. I chose 8 fsc (approx. 28.6MHz) for this project in order to decrease requirement to the post filter, like over sampling technic.

Wave form memory

The frame memory size is number of samples per line * number of lines per frame. In this case, memory size of 1820 * 525 = 1M samples is required for the frame memory, and the samples must be read out in transfer rate of 8fsc (cycle time of 35 nsec) continualy. An SDRAM is the best for shch use. However I have some small junk SRAMs (32 Kbyte) used for cashe memory, and I wanted to use these chips for the frame memory under recycle spirit :-)

To store the test pattern into the small memory, any data compression process is required. I had an eye to a point that the video test signal tend to repeats same line patterns. When store only the line patterns used in the frame and select the required line pattern for each line, the memory size and downloading time will able to be reduced drastically. Ten several line banks will do for the most test patterns except for patterns change vertically, such as picture, monoscope and vertical sweep. Luminance component and chrominance component are separated into two channels. The luma signal has many kind of patterns at vertical blanking area, the chroma signal is fliped every line. The effeciency of compression ratio can be improved when separated the NTSC signal into luma and chroma components. Y/C signal can be generated at the same time.

This compression technic can be applied to only NTSC format because the chroma subcarrier at the PAL system, fh*283.75+25 Hz, slips its SC-H phase in rate of 360 degrees per frame, each chroma line patterns in a frame are not the same. SECAM is out of the question. For these video formats, complete frame memory is needed. Don't ask me :-p

Controller

An Atmel ATmega161 is used as controller. It manages user interface, downloading the wave form data to the line memory and selecting wave form for each line under line sequense list. The line patterns are shrinked to 1/2-1/50 based on its monotony of luma and periodicity of chroma, The shrinked data is stored to a serial EEPROM and expanded at downloading, 59 frames (100 Mbytes) are packed into 40 kbytes. Therefore two compression process are applied and attained the total compression ratio of approx. 1/2500.

Analog Outputs

A three channel DAC is used. One channel is assigned for a luma channel, left two channels are for chroma channels. The reason of two DAC channels are assined to chroma is to support Y/CB/CR component output. Of course NTSC mode and component mode cannot be used at the same time. The DAC outputs are filtered, bufferred and then output. CR components placed at inverting input is to compensate frequency response due to aperture effect.

Contorl Panel

Two digits numeric LEDs indicate the frame pattern currently output and one of two video format incicater at the corresponding connector is lit. The frame pattern is selected with a dial and set by clicking the dial. Numeric LEDs blink during the pattern selection.

Video Line Selector

2009-05-07T01:03:00.000-07:00

When measure video signal with oscilloscope, video line selector is very useful to find a scan line. The line selecter generates trigger pulse at selected line, oscilloscope will display only selected line. This is a very simple video line selector.

Now, I have a Tektronix TDS3032B digital oscilloscope for home use. It can also be implemented video line selecter feature with a video module, any external accessory might not be needed. But this project has started before purchasing the new one, so that I achieved the project to open to the public.

Specifications:
Trigger mode All-line, Both-field, Odd-field, Even-field, An Odd-field in superframe, An Even-field in superframe
Control panel Display: 16cols x 1row LCD module
Conrol: UP button, DOWN button, MODE button
Interface Video input: CVBS/Y Input/Through (75ohms or Hi-Z)
Trigger out: TTL level (rise edge)
Power supply DC 5..12V, 20mA

HARDWARE

There are many video line selecter projects from of old, most line selecters are composed with a sync-separator and some counter ICs. In this project, all of counter/trigger function are processed by a microcontroller without external counter. Therefore, the circuit diagram could be very simple and it has many function.
Compsite sync pulses separated from input video signal are tied to external interrupt of an AVR and a flip-flop. Processing trigger output only software cannot avoid propagation delay and jitter, so that made the trigger edge pass through to the trigger output directly. Sync separator is composed by discrete parts, however, LM1881 is recommended if it is in stock. Test pins are for monitoring the clamped wave form.

SOFTWARE

Firmware for NTSC and PAL

Foreground task processes only user interface. Counting incoming syncs and trigger control are procecced by interrupt driven background tasks. AVR has very high performance, it will able to be implemented some additional functions. Any customized trigger mode or superimposing line marker will easy to implement with modifying the firmware or expanding some external components.

Line counting

First, 16 bit timer/counter is initialized as free running counter with 1.25MHz source clock and compare register is set to 65. The value will reach 79 while a horizontal period. The line counting process is driven by external interrupt (INT1). In this interrupt, when timer/counter has exceeded 60 (3/4H), the timer/counter is cleared and line counter is increased. If timer/counter is less than 60, the interrupt is half-H pulse, the line counter is not updated.

Trigger

After line counter is updated, if the value (next line) matches trigger line, trigger request flag is set. Compare match interrupt (timer/counter maches 65) is occured every 10µs before next line start. In this interrupt, if the trigger request flag is set, reset to external flip-flop is released and next sync edge will pass through the flip-flop. The flip-flop is reset agan by the external interrupt process. The trigger pulse width is approx. 2µs.

Detecting vertical sync pulse

8µs after external interrupt occured, sync level is sampled and stored it into shift register (left shifted). If the interrupt is at half-H, exit with no process. If it is start of line, compare the value of shift register and 0b11111110 (sync pattern at start of vertical sync). Only line 4 in odd field will match this condition. When the vertical sync is detected, set line conter to 4. This process is before updating line counter.

Detecting no signal

If no sync is detected for 20ms, timer/counter overflow interrupt will occure. In this interrupt, no signal flag is set and the condition is informed main task. The no signal flag is cleared by external interrupt (INT1).

GPIB to RS-232 converter

2009-05-05T01:44:00.000-07:00

This project fills the need of anybody who has a test instrument with the GPIB port and likes to get the screen dump on his PC without any GPIB card. It emulates the HP7470A operation on the GPIB side, and outputs the HP-GL data at the RS-232 port to be read and stored on the PC by any suitable software. The operation of this interface is not just limited to plotter emulation: any data intended to be received by a GPIB Device (addressable or listener only) can be captured and brought out to the RS-232 port, including raw data from the instrument or rasterized data for a GPIB graphic printer. GPIB addresses and other set-and-forget parameters can be modified and permanently stored using a simple setup menu. It is based on a PIC16F628A microcontroller, and the PCB size is just 7x7.5cm.

SCHEMATIC AND DETAILS

The hardware of Pic-Plot interface is quite simple: the active components are just a PIC16F628A, a 5V regulator and three transistors. External connections are a GPIB connector, a Serial port and a DC power connector.

The microcontroller does all the necessary jobs to emulate GPIB Device functionality, in both Listener and Talker mode, by recognizing addressing, commands and managing the Handshake lines. Controller mode is not needed for the intended functionality, and therefore is not supported. Once the device is addressed and it receives data from the Talker, the same data are forwarded to the COM port at 9600 baud: the hardware UART inside the PIC16F628A generates the serial data going to the PC through the RS232 port. Only in Setup mode the data flow is bidirectional at the same baud rate. The RS-232 connector on the interface is a standard male DB-9, and should be connected to the PC using a null-modem female-to-female serial cable.

PCs missing the COM port but equipped with an USB port can be still used with the aid of an inexpensive USB to serial converter, provided that the necessary Virtual Com Port drivers are properly installed.

A jumper is provided to enter Setup mode: when Pic-Plot it powered with this jumper in the closed (short) position, then it starts-up in Setup mode. In this mode the microcontroller UART is used to read/change a few set-and-forget parameters. GPIB cable can be left connected to the instrument, but in Setup mode the GPIB port is not monitored by the Pic-Plot. For normal operation this jumper must be left open. More details about Setup mode can be found in the USE AND OPERATION section.

Power supply can be any voltage between 8 and 16V, and current drain is far below 20mA. With such a large supply requirements, a low-cost unregulated 12V wall adaptor can be used as a power source. Connector polarity is center positive (+). An interesting possibility for those who use the USB-to-serial bridge is to bypass the Pic-Plot 5V onboard regulator and spill the 5V supply from the USB connector mounted on the bridge. This solution of course asks for a simple modification of the bridge or a modified USB cable, then it is suggested only to people having the necessary technical skills to do things right. You can find details by clicking here, or you might prefer to see our new Pic-plot2 which directly supports GPIB-USB conversion.

High Gain Amplifier

2009-04-28T21:39:00.000-07:00

The amp is based on the High Gain PCB, so uses a pair of LM3876 (or LM3886) power opamps, run from a ±35V supply. I used a cut-down P88 preamp PCB because I only wanted one preamplifier stage, but the entire board can also be used. Alternatively, the P19 amp can be run at higher gain than normal, alleviating the need for a preamp at all. The down side of this is that the noise level will be higher, and background noise may be audible with efficient speakers and/ or very quiet surroundings.

The internal layout can be seen best in Figures 2 and 3. The main heatsink runs down the middle of the amp, and it separates the input and output stages. The material is 10mm thick aluminium, 45mm high and 180mm long. Because this is a prototype of the chassis assembly, there are several things that I would do differently if I build another. The chassis is more complex than it should be, and there are several opportunities for simplification. These became obvious after the basic chassis was well underway (naturally), and there were holes that I couldn't 'undrill' to simplify construction. Such is life.

The front top view shows the general layout of the amp's internals. On the left is the sheet aluminium clamp that holds the capacitors in place, and against the central heatsink section is the P19 amp board. On the other side of the heatsink is the input selector switch and then the ½ P88 board.

Along the rear (from left to right) is the DC connector, speaker outputs and inputs. As it turns out, 4 inputs is enough for my application, and had I restricted it to that the shield between the last set of inputs and the speaker connectors would not have been needed.

The DC connector, speaker connectors and input RCA sockets are all mounted on blank fibreglass PCB material to insulate them from the chassis. Where needed, the copper was removed to create a rudimentary PCB pattern - this is evident on the DC and speaker panels. The boards were 'etched' using a rotary tool (Dremmel or similar). Although the resolution and accuracy are not good enough for an amplifier, this method works very well for such applications.

The back view shows the vent slots along the top, and you can see that the RCA connectors do not contact the chassis. Naturally, the speaker terminals are insulated. The DC connector is clearly visible on the right. It is a lot easier to simply make the back panel a little shorter than the other panels than it is to cut slots as shown. Even with a milling machine, these are somewhat tedious to do, and it is difficult to get perfect alignment without proper jigs. The hole for the DC plug and socket is relatively easily made using a drill and square file. The switch hole will require some fairly tedious filing if you use a rectangular switch as shown, however you can use any switch at all, because it only has to switch 9V AC.

Again, the slots look cool, but a series of holes will work just as well. There are a number of other refinements as well, and these are listed in the construction section below.

The Electronics

As noted above, the electronics are based on two existing projects - P19 stereo 50W amplifier, and P88 high quality preamp. The schematic is shown below (one channel only), and the P88 only uses the second half of the PCB. The P19 power amp is constructed normally, and there are no changes from the published project.

The inputs can be designated with whatever you want, and you can add more if desired (within the limits of the rear panel real estate). It is important that the gain of the preamp section is kept low enough to ensure that none of your inputs will clip the opamp. Assuming that CD/ DVD players are capable of about 2V, this means that the gain must be kept below 6.5 (16dB). This is not a problem unless you change the values of R7A, B and C, since the maximum gain is limited to about 9.5dB with the values shown.

The caps before and after the volume control can be bypassed completely (using wire links), but I do not recommend that you do so. If there is DC across the pot,it will become noisy and scratchy after a while. Even small amounts of DC can cause problems.

Power Supply Module

The power supply I used is probably overkill, but I simply used parts I had on hand. The schematic is shown below. Although I used zeners for the opamp supply as shown, some constructors are bound to be uncomfortable with such a simple arrangement. The P05 board can be used to provide full regulation, but with only one dual opamp, I'm not sure it is warranted.
A photo of the cpmplete module is shown below. The soft start isn't really needed with a 160VA transformer, but it does no harm, and allows remote low voltage switching. Since this was a requirement (the connectors are illegal for use with hazardous voltages), it was a small price to pay. Although the transformer is happy without the soft start, there is a total of 20,000uF on each supply rail, and this would place great stress on the bridge rectifier.

The two 2.2k 1W resistors across the filter caps in the supply box ensure that the caps will discharge even if the amplifier is not connected. They are not strictly needed, but are recommended to prevent nasty sparks is the amp is connected while the caps are still charged. Large electros can easily maintain a respectable charge for many hours.

The power supply is conventional in almost all respects. I used a 160VA transformer, a 400V 35A bridge rectifier, and a total of 20,000uF per supply rail - 4 x 10,000uF caps in all. When the connecting cable resistance is added in, there is almost no ripple at all at the amplifier, even with both channels at full power. The cable resistance aids filtering, but at the expense of slightly reduced maximum continuous power. I obtained over 40W per channel with both channels driven into an 8 ohm load, and peak short term power is over 60W / channel.

You can use less capacitance of course, but with some increase in ripple and (perhaps) noise. For an amp of this nature, I expect that few constructors will want to use less than about 4 x 4,700uFcaps. Additional capacitance can also be used in parallel with the zener diodes, but 100uF 16V caps fit the P88 board easiliy. There is nothing to suggest that more capacitance will serve any purpose.

Since the amplifier is absolutely dead quiet even at full volume with unterminated inputs, there is nothing one can do to make it any better. Placing one's ear right next to the speaker (one of average sensitivity), circuit noise is just audible. There is no hum at all.

Super Amplifiers 300W Output Power

2009-04-26T18:42:00.000-07:00

The circuit described on this page is a modification of the original Double Barreled Amplifier. The circuit has been simplified somewhat. The circuit board layout is smaller and much more compact. The driver transistors now mount on the circuit board instead of on external heat sinks. And the circuit has the feedforward compensation that I describe for the Low TIM Amplifier.

The original circuit board for one channel had eight 5-watt resistors on it, one in series with the emitter of each output transistor. On the new layout, four of these have been moved to the heat sink channel where they solder between pins of the transistor sockets. This change not only helps make the circuit board smaller, but it eliminates eight wires between the heat sink and the circuit board. One of the figures below illustrates how these resistors are installed in the heat sink channel.

If you build this amplifier, you must keep the wiring between the heat sinks and the circuit boards as short as possible if you don't want oscillation problems.

When you test the circuit boards before connecting the power transistors, temporarily connect a 10 ohm resistor in series with a 0.1 ufd capacitor from the loudspeaker output to the power supply ground.

The Circuit Boards

We do not have circuit boards for the Double Barrelled Amplifier. If you wish to build it, you must make your own. Two drawings show the parts layout on the board, one with circuit traces and one without. These are scaled by a factor of 1.5. The other shows the circuit traces only. All layout views are from the component side of the board. You must flip the layout for the foil traces over to obtain the solder side view. The circuit board measures 4 inches by 6 inches. To my knowledge, there are no errors in the layout. If you decide to use it, you should carefully check it for errors because I could have easily made a mistake.

We do not recommend that you make the circuit boards unless you have experience in doing it. A source of materials for making your own printed circuits can be found here. I have been told that their "Press and Peel Blue" product (not the wet stuff they sell) can be used to successfully make boards with traces as narrow as 0.01 inch. The smallest traces on the amplifier layout are 0.03 inch wide. The PnP Blue product is basically a transfer medium that allows you to transfer the toner image from a laser printer directly onto bare copper clad board and then etch it in FeCl3 (ferric chloride).

After you etch the board, the copper should be cleaned with steel wool, lightly coated with solder flux, and then "tinned" with a soldering iron and rosin core solder. Do not use a commercial tinning solution that you dip the board into. It is almost impossible to solder a board that is tinned with one of these products because they corrode very quickly. When you drill the board, you should use the correct size drill bit for the pads. The hole diameters I recommend are: small pads - 0.032 inch, medium pads - 0.040 inch, large pads - 0.059 inch, mounting holes - 0.125 inch. If you do not use a sharp drill bit, you can pull the pads off the board when you drill it.

Circuit Description

If you compare the Double Barreled circuit to the Low TIM circuit, you will see a lot of similarity between the two. Indeed, there is a Low TIM Amplifier embedded in the Double Barreled Amplifier. The major difference between the two is that transistors are added in series with those in the Low TIM circuit to form the Double Barreled circuit. By doing this, the voltage across the transistors is decreased so that the power supply voltage can be increased for higher output power.

Basically, the circuit description for the Low TIM Amplifier also applies to the Double Barreled Amplifier. The major difference between the two is the addition of transistors Q22 through Q31. Q22 is connected as a common base stage at the output of Q12. The two transistors form a cascode stage. The base of Q22 connects to the junction of R52 and R54. These two resistors are equal and are connected as a voltage divider between the loudspeaker output and the positive rail. This forces the base voltage of Q22 to float half way between the loudspeaker output voltage and the positive power supply rail. Similarly, Q13 and Q23 form a cascode stage. R53 and R55 force the base of Q23 to float half way between the loudspeaker output voltage and the negative power supply rail. The addition of Q22 and Q23 cause the collector to emitter voltages of Q12 and Q13 to be approximately one-half of what the voltages would be without Q22 and Q23.

Transistors Q24 and Q25 connect in series with the pre-driver transistors Q14 and Q15. The base of Q24 floats half way between the output voltage and the positive rail. The base of Q25 floats half way between the output voltage and the negative rail. The addition of Q24 and Q25 cause the voltages across Q14 and Q15 to be approximately one-half of what they would be without Q24 and Q25. Similarly, transistors Q26 through Q31 cause the voltages across Q16 through Q21 to be approximately one-half of what they would be without Q26 through Q31. By connecting the transistors in series in this way, the rail voltages can be increased for higher output power.

The basic construction details of the Low TIM Amplifier also apply to the Double Barreled Amplifier. There are two short circuit jumper wires that must be soldered on the circuit board. These are marked with a J on the layout. In addition, you must solder a short circuit jumper in place of C6B if you use a non-polar capacitor for C6A. This is explained in the parts list for the Low TIM Amplifier. Because there are eight output transistors, two main heat sinks per channel are required. Q18, Q20, Q28, and Q30 should be mounted on one and Q19, Q21, Q29, and Q31 on the other. Resistors R61 through R64 and wires connecting the collectors of Q18 and Q20 and the collectors of Q19 and Q21 mount on the heat sinks. These connect between lugs on the transistor sockets. The four bias diodes D1 through D4 can be mounted on either heat sink. It is not necessary to divide the diodes between the two heat sinks because both heat sinks will operate at the same temperature. I recommend setting the voltage across Q7, i.e. the voltage between the collectors of Q22 and Q23, so that that amplifier is biased at 120 mA. This will give the same quiescent power dissipation per heat sink as in the Low TIM Amplifier.

Testing the Circuit Boards

After you solder the parts to the circuit board, it is tested using the same procedure specified for the Low TIM circuit board. First, you must solder the short circuit jumper across Q7 and you must solder the 100 ohm 1/4 W resistors from the loudspeaker output to the emitters of Q16 and Q17. If you don't have a bench power supply that puts out plus and minus 85 to 93 V dc, you can test the circuit board at a lower voltage. I would prefer test voltages of at least plus and minus 50 V dc. An option is to connect bench power supplies in series to obtain the plus and minus 85 to 93 V dc. I have routinely connected two 40 V Hewlett Packard power supplies in series with the positive and negative outputs of a Hewlett Packard 50 V dual power supply, and I have never had any problems. To protect the circuit boards, you might want to put a 100 ohm 1/4 W resistor in series with the plus and minus power supply leads for the tests. The current drawn by the circuit should be low enough so that the voltage drop across these resistors is less than 1 V if nothing is wrong on the circuit board. There are 2 ground wires from the circuit board. Both must be connected when testing the boards.

I can't stress how important it is to be careful in testing a circuit board. Even simple errors can cause the loss of many expensive transistors. I always use current limited bench power supplies to test a circuit board before and after connecting the power transistors. I also bias an amplifier using current limited power supplies in place of the amplifier power supply. When I initially power up an amplifier with its own power supply, I always use a Variac variable transformer to slowly increase the ac input voltage from 0 to 120 V rms while observing the amplifier output on an oscilloscope with a sine wave input signal. If I see anything wrong on the oscilloscope, I turn the Variac to zero and try to diagnose the problem using the bench power supply. I never use a load on the amplifier for these tests.

Parts List

With the following exceptions, the parts for the Double Barreled Amplifier are the same as for the Low TIM Amplifier.

Capacitors

C10, C11 - 15 pF mica
C13, C14 - 100 uFd 100 V radial electrolytic
C21, C22 - 47 uFd 100 V radial electrolytic
C26, C27 - 270 pF mica
C28 - 0.01 uFd 250 V film

Transistors

Q1, Q2, Q5, Q7, Q9, Q10 - MPS8099 or MPSA06
Q3, Q4, Q6, Q8, Q11 - MPS8599 or MPSA56
Q23, Q24 - 2N3439
Q22, Q25 - 2N5415
Q26 - MJE15030
Q27 - MJE15031
Q28, Q30 - MJ15003
Q29, Q31 - MJ15004

Diodes

D5, D6 - 1N4934 fast recovery rectifier
D13 through D16 - 1N5250B 20 volt zener diode

Resistors

R13, R14 - 5.6 kohm 1 watt (This value is for 85 V power supplies. For other power supply voltages, the formula is on the Parts List page for the Leach Amp.)
R28, R29 - 200 ohm 1/4 watt
R30, R31 - 3.9 kohm 1 watt
R37 through R40 - 470 ohm 1/4 watt
R41 through R44 - 10 ohm 1/2 watt (changed 6/27/00)
R52 through R55 - 6.2 kohm 1 watt
R56 through R59 - 10 ohm 1/2 watt (changed 6/27/00)
R60 - 39 ohm 1/4 watt
R61 through R64 - 0.33 ohm 5 watt. These 4 resistors are mounted on the heat sinks between solder lugs on the power transistor sockets. The wires that connect the collectors of Q18 and Q20 and the collectors of Q19 and Q21 are also soldered between the lugs on the sockets. Keep all leads as short as possible and use insulation stripped from hookup wire around the bare leads of the resistors.
R65, R66 - 300 ohm 1/4 watt

Heat Sinks

Double the number of heat sinks required for the Low TIM Amplifier.

Power Supply Components

The power supply circuit diagram is the same as for the Low TIM Amplifier. The parts are the same with the following exceptions.

T1 - The transformer should have either a center tapped secondary or two separate secondary windings which can be wired in series. With 120 V ac rms applied to the primary, the no load secondary voltage should be 120 to 130 V ac rms for a center tapped secondary or 60+60 (60x2) to 65+65 (65x2) V ac rms for two secondary windings. This should give a no load amplifier power supply voltage of plus and minus 85 to 93 V dc. Some transformers are rated at 115 V ac rms on the primary. With 120 V ac rms applied, the secondary voltage will be greater by a factor 120/115. If the transformer is rated at full load, its no load voltage will be 15% to 20% higher. I would recommend a transformer current rating of at least 6 A. The transformer I used in each of my two original Double Barreled Amplifiers was the Signal 230-6. It had two center tapped 115 V 6 A secondaries which I wired in parallel to obtain a secondary rating of 115 V at 12 A. The primary had three voltage taps: 105 V, 115 V, and 125 V. I wired the AC line input to the 115 V tap. With 120 V AC applied to the 115 V tap, I got plus and minus 85 V DC on the power supplies and 270 W into an 8 ohm load. If I had used the 105 V primary taps, the power supply voltage would have increased to about 93 V and the amplifiers would have put out over 300 W. The Signal transformer was definately an overkill. It weighed 38 pounds. But it would really kick you know what. To my knowledge, this transformer now is available only by special order.
C1P, C2P - I used two Mallory CG832U100G1 8,600 uFd 100 V capacitors in parallel for each of these so that I had 34,400 uFd total in each of my two amplifiers. This was probably an overkill. The energy stored in the eight apacitors was about 250 joules. This is enough energy to lift a 25 pound dog over 7 feet off the floor. For C1P and C2P, I would recommend at least 10,000 uFd total for each. The voltage rating should be 100 V or greater.

New Vacuum Tube Amplifier

2009-04-26T17:48:00.000-07:00

Features

Output: 4-6550's in triode-mode class AB2 push-pull parallel. About 80 watts RMS per channel.
No global negative feedback. Several local loops with limited negative feedback.
Ultra-wide bandwidth Plitron toroidal output transformer.
Servo to maintain precise dc-balance in the output circuit.
MOSFET-regulated power supplies. Relative rather than absolute voltage reference.
Designed using extensive PSpice computer simulation.
Constructed as a pair of monoblocks.

Schematic diagram

Notes on the schematic diagram

The power supply has been simplified-- Power transformers and rectifiers have been omitted and some parts have been omitted from the MOSFET voltage regulator circuits: 1N5242 zener diodes between the source and gate and 10k resistors in series with the gate. These parts serve as protection in case of accidental short circuits, but don't affect the operating point. The full power supply schematic is shown below.
6SN7's are used instead of 12AU7's for the driver tubes. They have the same plate characteristics, but they have higher maximum plate voltage (450 vs. 330 V) and greater plate dissipation (3.75 vs. 2.75 W per section). Think of the octal-based 6SN7 as a 12AU7 on steroids.
The NODESET blocks (lower right) initialize dc levels in the bias servo so the simulation runs properly. They don't exist as physical entities.
All resistors are 1/2 watt unless noted, and with the following exceptions. R3P, R4P and R5C through R8C (68K) are 2W. R9C through R12C (20 ohm) are 1W: I used 2-10 ohm resistors in series to make them. R9S through R12S (200 ohms) are 2W.
Capacitor voltage ratings: It never hurts to go over the minimum, though the capacitors will be larger and may cost more. I often use caps with higher voltage ratings because I have them on hand or found them at a good price in an electronics surplus shop. Here are some minimum ratings: C1G: 100V. C2G: 400V (600V would be better; necessary without the time delay); C3G and C4G: 400V (600V would be better); C3M and C4M: 400V; C5G through C8G: 600V; CBS2, CBS4, CBS6 and CBX1 through CBX3: 100V. The voltage ratings of many of the power supply capacitors are shown on the circuit board wiring diagrams, below.
Some of the feedback connections may be a little hard to trace. ORN goes between the 20 ohm output transformer secondary feedback winding and R3F near TU3. Similarly, VLT goes to R4F near TU4. BLU and BLK on the output transformer secondary speaker winding go to the output tube cathode circuits.
RLS (5 ohms) is a simulated speaker load.
CRF: All three nodes are connected together.
BRN and VIO are not used. They are the ultra-linear taps. In the original version of TENA there was a switch to select output tube screen grid connections between BRN and VIO (UL mode) and GRN and YEL (triode mode).

Input stage/phase inverter

Input stage TU1 is a simple voltage gain stage with local negative feedback, derived from the R1B, R1C voltage divider. It is capacitively coupled to split load phase inverter TU2. The capacitor has an unusually low value-- 0.01 µF-- because TU2 has an exceptionally high input impedance-- several Megohms. The advantage of capacitive coupling is that it allows the voltage level in TU2 to be set for maximum output and it allows the ac current in TU2 to be precisely equal to, but 180 degrees out of phase with, the current in TU1. The net ac current drawn by these two tubes from V+420 is therefore zero. This is an effective way of isolating the audio signal from the power supply, which doesn't need to supply ac current. In conventional designs ac signal often has to flow through electrolytic capacitors, which are grungy leaky devices with memory-- harmful to audio quality. I designed TENA to draw zero net ac current from all power supply outputs (easy to do in a push-pull design), at least up to the power level where one of the output tube pairs starts cutting off.

Toroidal output transformer

We chose the Plitron toroidal transformer because of its exceptional bandwidth: -3 dB at over 200 kHz, the result of high primary inductance (the good stuff) and low leakage inductance (the bad stuff-- kind of like HDL and LDL cholesterol)-- much better than can be achieved with a conventional EI transformer. High bandwidth is important because output transformers have an intrinsic second order rolloff, which can make them unstable in the presence of negative feedback unless careful phase compensation is applied (see Feedback and Fidelity). Phase compensation reduces the bandwidth, which is not a problem with the Plitron toroidal transformers. But this bandwidth comes at a price-- toroidal transformers are much less tolerant of dc-imbalance than EI transformers; they may saturate at dc imbalances as low as 8 mA. (I don't know the exact number; I never simulated it.) You would have to set the bias of each tube individually, and then you'd have to worry about how the tubes age. So I designed a bias servo circuit to maintain perfect dc-balance under all circumstances except outright tube failure.

The Plitron PAT 4006CFB 100 Watt toroidal output transformer is not currently listed on Plitron's website, but I've heard (June 2003) that it is available. Contact Norman Woo. The closest models are the 4006, which lacks the special feedback winding, and the 2100-CFB which has a higher primary impedance. The minimum feedback version of TENA (below) works with the 4006.

Bias servo and adjustment

The time-averaged (low pass filtered) dc current of an output tube operating in class AB fixed bias is relatively constant at low power levels but increases at high power levels. For this reason a fixed voltage cannot be used as a reference for biasing the output tubes. One tube (TU9, driven by TU5) operates at fixed bias, and its low pass filtered cathode voltage (CRF) is used as the reference for biasing the other tubes.

The bias servo is illustrated in the lower left of the schematic. It uses the LM324 quad op amp-- cheap but perfectly adequate. Inputs U1A, U1B and U1C of the LM324 compare cathode voltages 10C, 11C, and 12C with reference voltage CRF, which is the voltage on cathode 9C low pass filtered with RBS2 = 33k and CBS1 = 10µF ( located near U1B on the schematic). The LM324 outputs control the P-channel MOSFETs, each of which controls a voltage divider between VBB (-90V) and VOP (+12.5V) to deliver the appropriate bias voltage to the driver grid circuits (BIAS_6, BIAS_7, and BIAS_8). This measures between -45 and -50V in my amplifiers, which operate at 60 mA plate current. Audio purists please note: the servo operates at extremely low frequencies; the op amp and MOSFETs are well outside the audio signal path.

A single potentiometer, RB5 (in the VBB supply, bottom center), controls the bias current directly in TU9, and all the other tubes indirectly through the servo. Bias current may be measured across any of the 20 ohm resistors R9C-R12C as E/20. They should all be the same if the servo is working properly. 1 to 1.2 volts is a good nominal value, corresponding to 50 to 60 mA per tube (70 mA was used in the Dynaco Mark III). Increasing the current increases power consumption and reduces tube life and output power, but moves you closer to Class A (where both tubes always conduct).

Class AB2 output stage and drivers

Class AB2 differs from the more common class AB1 in that the output stage grid is driven positive-- it draws grid current-- at high power levels. Class AB2 has no advantage for output tubes operating in pentode mode and little advantage for ultra-linear mode. But it results in a huge power boost for output tubes operating in triode mode. You can get almost as much power out of class AB2 triodes as you can out of class AB1 pentodes.

If you try to do operate in class AB2 with conventional capacitive coupling, the coupling capacitor starts charging as soon as grid current is drawn. This drives the grid negative-- toward cutoff, and it recovers with the RC time constant of the coupling capacitor and grid resistor. To operate successfully in class AB2, the output stage must be either transformer or direct coupled. I chose direct coupling because interstage transformers are expensive and have limited bandwidth.

The direct coupled drivers are the source of much of TENA's complexity. Because the quiescent grid voltage of each output tube must be set individually to control its quiescent (dc) current, one driver tube (TU5-TU8) is required for each output tube (TU9-TU12). Cathode followers (CF's) were chosen because they have low output impedance and can source the needed output tube grid current. The cathodes have to be somewhere near -50V to properly bias the output tubes. This means the CF must be driven by voltages outside the range of conventional power supplies, hence the need for VDR- and VDR+: the price of perfection. In reviewing the design I find that the driver tubes may be operating a little too conservatively-- dissipating only 0.78 W (of a 6SN7 maximum of 3.75 W). I've discussed driver dissipation under PSpice output, below. I may increase VDR+ from 205 to around 250 V by increasing RD1 from 470k to 680k. This would reduce the power dissipation in MOSFET MD1.

Output tube grid stop resistors R9G-R12G play an important role in TENA's soft clipping. When power levels become high enough level for grid current to be drawn, a voltage drop across these resistors gradually limits the plate current. Soft clipping consists of low order harmonics which have much less adverse effect on sound quality that the high order harmonics characteristic of abrupt clipping. But total harmonic distortion for soft clipping amplifiers tends to be higher. Yes, lower harmonic distortion doesn't mean better sound. See "The great harmonic distortion scam" in Feedback and Fidelity. TENA oscillated when the grid stop resistors were removed. This was the only performance feature PSpice didn't catch. The reason is that the output transformer model is somewhat simplified-- it's extremely difficult to model its distributed capacitance.

Power supplies

The time delay circuit (U3 (the 555B chip), Q1, Relay_SPDT_nb, RT1, CT1, CT2, RT3, D1, RV1, and RT4) has apparently never been implemented. RT4 should be replaced by a straight wire; VBIN is connected directly to NTC (negative temperature coefficient; 50 ohms cold; Mouser527-3504-50) thermistor RV10.

The precise values of most of the capacitors in the power supply, particularly CV1, CV2, CB1, CB2, CD1 and CD2, are not critical. In many cases they were determined by parts availability. If the values are 2 uF or under they are film capacitors. If they are over 2 uF they are electrolytics.

Depending how you count there are two (power transformers), four (rectifier circuits) or six (voltage levels). All use fast recovery rectifier diodes. All except VDR- are taken from the mighty Plitron 854710 toroidal power transformer, which I can't seem to find in their catalog. Toroidal power transformers perform well, but they have less of an advantage than toroidal output transformers-- you don't need wide bandwidth for 60 Hz. The CL80 inrush current limiter limits turn-on current in the tube filaments.

Universal Input Linear Fluorescent Ballast

2009-04-25T01:31:00.000-07:00

Features

Drives one 35 W TL5 Lamp
Input Voltage: 80 VAC to 260 VAC
High Power Factor/Low THD
High Frequency Operation
Lamp Filament Preheating
Lamp Fault Protection with Auto-Restart
Low AC Line Protection
End of Lamp Life Shutdown
IRS2166D(S)PbF HVIC Ballast Controller

The Board is a high efficiency, high power factor, fixed output electronic ballast designed for driving rapid start fluorescent lamp types. The design contains an EMI filter, active power factor correction and a ballast control circuit using the IRS2166D(S)PbF Ballast Control IC1.

The Board consists of an EMI filter, an active power factor correction section, a ballast control section and a resonant lamp output stage. The active power factor correction section is a boost converter operating in critical conduction mode, free-running frequency mode. The ballast control section provides frequency modulation control of a traditional RCL lamp resonant output circuit and is easily adaptable to a wide variety of lamp types. The ballast control section also provides the necessary circuitry to perform lamp fault detection, shutdown and auto-restart.

This board is designed for single TL5/35W Lamp, voltage mode heating (JV1 and JV2 mounted, JC1 and JC2 not mounted). TL5 lamps are becoming more popular due to their lower profile and higher lumen/ watt output. These lamps, however, can be more difficult to control due to their higher ignition and running voltages. A typical ballast output stage using current-mode filament heating (filament placed inside L-C tank) will result in excessive filament current during running. The output stage has therefore been configured for voltage-mode filament heating using secondary windings off of the resonant inductor LRES. The lamp has been placed outside the under-damped resonant circuit loop, which consist of LRES and CRES. The filament heating during preheat can be adjusted with the capacitors CH1 and CH2. The result is a more flexible ballast output stage necessary for fulfilling the lamp requirements. The DC blocking capacitor, CDC, is also placed outside the under-damped resonant circuit loop such that it does not influence the natural resonance frequency of LRES and CRES. The snubber capacitor, CSNUB, serves as charge pump for supplying the IRS2166D.

The IRS2166D Ballast Control IC is used to program the ballast operating points and protect the ballast against conditions such as lamp strike failures, low DC bus, thermal overload or lamp failure during normal operations. It is also used to regulate the DC bus and for power factor control allowing high power factor and low harmonic distortion.

Switch Mode Power Supply 100W - 16 at 7A

2009-04-25T00:48:00.000-07:00

Generally, Schottky diodes are traditional devices use in passive rectification in order to have low conduction loss in secondary side for switching power supplies. The proliferations of synchronous rectification (SR) idea - which is mostly use in buck-derive topologies - have reached the domain of flyback application in recent years. The use of low-voltage-low-Rdson mosfet has become so attractive to replace the Schottky rectifiers in high current applications because it offers several system advantages such as dramatic decrease in conduction loss and better thermal management of the whole system by reducing the cost investment in heat sink and PCB space.

A number of techniques in the implementation of SR in flyback converters are continuously growing from a simple self-driven (secondary winding voltage detection) to a more complex solution using “current transformer sensing” or combinations of both to improve the existing technology. The idea has become quite complicated though and additional discrete devices have made the cost and part counts issue even worse. Moreover, the issue of reverse current conduction (-due to the delay in sensing the sharp drop of secondary current during turn-off phase of the SR) still lingers on in different input line/ output load conditions. The use of a simple fast-rate-direct-sensing of voltage drop across the mosfet (Vsd) using integrated solution has pave the way for a much simpler and effective means of controlling the SR mosfets as well as alleviating the reverse current and multiple-pulse gate turn-ON issues.

The board is a universal-input flyback converter with single DC output capable of delivering continuous 100W (@ +16V x 6.25A) during active rectification mode. This board is primarily designed to study synchronous rectification using IR1166 in low-side configuration to take advantage of simpler derivation of Vcc supply from converter’s output. It is equipped with necessary jumpers to ease exploring the conduction behavior of synchronous rectifiers SRs in quasi-resonant mode, so discussion would be confined to variable frequency switching in Critical Conduction Mode.

It features the fast Vsd sensing of the IR1166 Smart Rectifier Control IC with gate output drive capability of 1.5Apk. It drives 2 pcs. of SRs in parallel (100V N-ch mosfet IRF7853 in SO-8 package with very low Rdson in its class : 18 mΩ max). This had greatly simplified the overall mechanical design for not having those bulky and heavy heat sinks normally seen in high current flyback design using passive rectification.

CIRCUIT DESCRIPTION

The PCB design is basically optimized as a test platform to evaluate of active rectification using Smart synchronous rectification and as well as basic features of flyback converter operating in quasi-resonant mode.

This board has 2-pin connector ( CON1 ) for AC input and a time-lag type 3.5A fuse for input current overload protection. Minimum input filtering is provided (Cp1-Xcap) before AC input voltage (90-264VAC) is routed to a 6Amp-bridge rectifier (DB1).

Primary side controller (U2) basically drives the primary Mosfet Q1 to operate in Critical-Conduction mode to eliminate turn-ON switching loss thru ZVS (zero voltage switching only occurs when NVsec > Vdcin ) or thru LVS ( low-voltage switching when nVsec< Vdcin) to reduce capacitive losses of Q1 especially at high line condition. The switching frequency Fsw at full load varies from ~38 to ~76kHz typically from low to high input condition and falls back to minimum value (fixed ~ 6 -10kHz) to reduce input power during light load condition.

Auxiliary winding is loosely monitored by demagnetization pin4 of U2 through Dp3, Rp5 and Rp11 network that sets the OVP limit with Rp6 and Rp11 sets the over power limit of the converter.

Resonant capacitor Cp7 is added to augment the overall parasitic winding capacitance and the primary mosfet Q1’s Coss to achieve ZVS and LVS at low and high input line condition respectively.

Optocoupler U3 provides isolated output voltage feedback to the primary side. The output voltage level across load connector CON2 (+16Vo) is monitored and regulated by the V/I Secondary error amplifier U4 (AQ105 or AS4305) that also manages the output current limiting function by monitoring the voltage across the RS25-26 current sense resistors.

The power stage of the secondary is using 2-SO8 low IRF7853 synch-fets (SR) in parallel to implement the low-side synchronous rectification. In this configuration, it is simpler to derive the Vcc supply for the U1 (IR1166 SO8-IC) controller directly from the DC output Vout. Jumper J5 is used to isolate U1’s Vcc from Vout so that user may easily evaluate IC’s power consumption especially during standby load condition. In the absence of a sensitive low current probe, the quiescent current Icc through Dp4 can be calculated from the differential voltage across the Rs17. The decoupling capacitor Cs17 and Cs18 provides additional filtering which is necessary to clean high frequency noise especially when U1 is driving several mosfets (SR1 // SR2) with high Qg parameters normally associated with high currentlow voltage mosfets.

The Vd and Vs sense pins monitor the voltage (Vsd) across the sync rect mosfets and proper attention was taken during PCB routing to ensure the integrity of differential voltage Vsd. This is done by directly taking the signal Vd from the drain pins of SR1//SR2 using a dedicated trace.

Probe points as well as redundant test hook points are provided to facilitate easy probing of essential test waveforms.

Switch Mode Power Supply 12V 8A

2009-04-19T22:56:00.000-07:00

The UCC3807 family of high speed, low power integrated circuits contains all of the control and drive circuitry required for off-line and dc-to-dc fixed frequency current mode switching power supplies with minimal external parts count.

Click on the picture to enlarged

These devices are similar to the UCC3800 family, but with the added feature of a user programmable maximum duty cycle. Oscillator frequency and maximum duty cycle are programmed with two resistors and a capacitor. The UCC3807 family also features internal full cycle soft start and internal leading edge blanking of the current sense input.

The UCC3807 family offers a variety of package options, temperature range options, and choice of critical voltage levels. The family has UVLO thresholds and hysteresis levels for off-line and battery powered systems. Thresholds are shown in the table below.

The circuit shown above illustrates the use of the UCC3807 in a typical 100-W, 200-kHz, universal input forward converter produces a regulated 12VDC at 8 Amps. The programmable maximum duty cycle of the UCC3807 allows operation down to 80VRMS and up to 265VRMS with a simple RCD clamp to limit the MOSFET voltage and provide core reset. In this application the maximum duty cycle is set to about 65%. Another feature of the design is the use of a flyback winding on the output filter choke for both bootstrapping and voltage regulation. This method of loop closure eliminates the optocoupler and secondary side regulator, common to most off-line designs, while providing good line and load regulation.

DC Servo Amplifier with Negative Feedback

2009-04-18T03:30:00.000-07:00

And negative feedback amplifier voltage than current negative feedback amplifier as excellent transient nonlinear distortion and intermodulation distortion characteristics, the frequency amplifier curve flat, high and low frequency response more exhibition wide; more important is that circuit Will load impedance into the feedback network, it can change the speakers of such fierce resistance to the load compensation, coupled with stable and reliable performance, than the negative feedback voltage amplifier has more advantages, the current negative feedback current amplifier is widely For the modern high-fidelity audio amplifier.

Circuit above is an excellent performance, improve the design of the fever-100 W × 2 DC Servo Amplifier current negative feedback stereo amplifier, formed by the two TDA7294, the frequency response of 10 Hz ~ 100kHz. The use of sophisticated audio Yun-double as the two-channel amplifier DC Servo Amplifier output. Speakers from the protection of ASIC μPC1237HA driver completed the relay switch-mute and amplifier output DC offset protection, and other speakers. When the AC power plug, the relay will be delayed for some time speakers access amplifier; disconnecting the AC power when, μPC1237HA detected exchange loss, immediately disconnect the speaker to relay, the amplifier is the complete elimination of the set, Shutting down the transition process the impact of noise on the speakers.

In actual use, taking into account the electricity grid fluctuations Rectifier amplifier output voltage ± Vs volatile, in order to avoid over-voltage and high temperature in the state of damage TDA7294 (Note pressure in the temperature of 25 ℃ under the conditions, if the temperature exceeds 25 ℃, TDA7294 the value will subsequently reduce the pressure), the exchange recommended power supply voltage transformer CT-AC26V × 2.

Motor current control circuit with external power transistors

2009-04-18T02:34:00.000-07:00

The L165 is a monolithic integrated circuit in Pentawatt ® package, intended for use as power operational amplifier in a wide range of applications, including servo amplifiers and power supplies. The high gain and high output power capability provide superiore performance wherever an operational amplifier/power booster combination is required.

1.2W AUDIO AMPLIFIER

2009-04-18T01:45:00.000-07:00

A very useful audio amp in an 8-pin DIL package. The IC features a very low minimum working supply voltage of 3V, low quiescent current, good ripple rejection, no crossover distortion and low power dissipation. Maximum supply voltages is 16 Volts into 16 Ohms speaker, 12Volts into 8 Ohms and 9Volts into 4 Ohms.

The TBA820M is a monolithic integrated audio amplifier in a 8 lead dual in-line plastic package. It is intended for use as low frequency class B power amplifier with wide range of supply voltage: 3 to 16V, in portable radios, cassette recorders and players etc. Main features are: minimum working supply voltage of 3V, low quiescent current, low number of external components, good ripple rejection, no cross-over distortion, low power dissipation.

Output power: Po = 2Wat 12V/8W, 1.6W at 9V/4W
and 1.2W at 9V/8W.

8 Watts Audio Amplifier

2009-04-18T01:22:00.000-07:00

Nice small audio amplifier which use only few parts to give good quality sound. This amp can be used as a simple booster, the heart of a more complicated amplifier or used as a guitar amp. Although not perfect, this amplifier does have a wide frequency response, low harmonic distortion about 1.5%, and is capable of driving an 8 ohm speaker to output levels of around 8 watts with slightly higher distortion. Any power supply in the range 12 to 18 Volts DC may be used.

The TDA 2003 has improved performance with the same pin configuration as the TDA 2002.
The additional features of TDA 2002, very low number of external components, ease of assembly, space and cost saving, are maintained. The device provides a high output current capability (up to 3.5A) very low harmonic and cross-over distortion.

Completely safe operation is guaranteed due to protection against DC and AC short circuit between all pins and ground, thermal over-range, load dump voltage surge up to 40V and fortuitous open ground.

Water Softener Circuit

2009-04-18T00:57:00.000-07:00

That circuit is based at a technique to remove or neutralize the salt in water, and protect the pipes at home as well as the washing machines or our selves from salt. Its called water softener and its automated circuit using two 555 timers. The cost of parts is nearly 10$ and its very easy to build it.

HI-FI CAR-RADIO Amplifier

2009-04-18T00:00:00.000-07:00

Constraints of implementing high power solutions are the power dissipation and the size of the power supply. These are both due to the low efficiency of conventional AB class amplifier approaches.

Here above is described a circuit proposal for a high efficiency amplifier which can be adopted for both HI-FI and CAR-RADIO applications. The TDA7294 is a monolithic MOS power amplifier which can be operated at 80V supply voltage (100V with no signal applied) while delivering output currents up to ±10 A.

This allows the use of this device as a very high power amplifier (up to 180W as peak power with T.H.D.=10 % and Rl = 4 Ohm); the only drawback is the power dissipation, hardly manageable in the above power range. Figure 20 shows the power dissipation versus output power curve for a class AB amplifier, compared with a high efficiency one. In order to dimension the heatsink (and the power supply), a generally used average output power value is one tenth of the maximum output power at T.H.D.=10 %.

From figure below, where the maximum power is around 200 W, we get an average of 20 W, in this condition, for a class AB amplifier the average power dissipation is equal to 65 W. The typical junction-to-case thermal resistance of the TDA7294 is 1 oC/W (max= 1.5 oC/W). To avoid that, in worst case conditions, the chip temperature exceedes 150 oC, the thermal resistance of the heatsink must be 0.038 oC/W (@ max ambient temperature of 50 oC). As the above value is pratically unreachable; a high efficiency system is needed in those cases where the continuous RMS output power is higher than 50-60 W.

The TDA7294 was designed to work also in higher efficiency way. For this reason there are four power supply pins: two intended for the signal part and two for the power part. T1 and T2 are two power transistors that only operate when the output power reaches a certain threshold (e.g. 20 W). If the output power increases, these transistors are switched on during the portion of the signal where more output voltage swing is needed, thus ”bootstrapping” the power supply pins (#13 and #15). The current generators formed by T4, T7, zener diodes Z1,Z2 and resistors R7, R8 define the minimum drop across the power MOS transistors of the TDA7294. L1, L2, L3 and the snubbers C9, R1 and C10, R2 stabilize the loops formed by the ”bootstrap” circuits and the output stage of the TDA7294.

In figures above, the performances of the system in terms of distortion and output power at various frequencies (measured on PCB shown in fig. 19) are displayed. The output power that the TDA7294 in high efficiency application is able to supply at
Vs = +40V/+20V/-20V/ -40V; f =1 KHz is:
- Pout = 150 W@ T.H.D.=10 % with Rl= 4 Ohm
- Pout = 120 W@ ” = 1% ” ” ”
- Pout = 100 W@ ” =10% with Rl= 8 Ohm
- Pout = 80 W @ ” = 1% ” ” ”

Results from efficiency measurements (4 and 8 Ohm loads, Vs = ±40V) are shown by figures 23
and 24. We have 3 curves: total power dissipation, power dissipation of the TDA7294 and power dissipation of the darlingtons. By considering again a maximum average output power (music signal) of 20W, in case of the high efficiency application, the thermal resistance value needed from the heatsink is 2.2oC/W (Vs =±40 V and Rl= 4 Ohm).
All components (TDA7294 and power transistors T1 and T2) can be placed on a 1.5oC/W heatsink, with the power darlingtons electrically insulated from the heatsink. Since the total power dissipation is less than that of a usual class AB amplifier, additional cost savings can be obtained while optimizing the power supply, even with a high headroom.

Automatic Room Lights

2009-04-08T00:17:00.000-07:00

n ordinary automatic room power control circuit has only one light sensor. So when a person enters the room it gets one pulse and the lights come ‘on.’ When the person goes out it gets another pulse and the lights go ‘off.’ But what happens when two persons enter the room, one after the other? It gets two pulses and the lights remain in ‘off’ state. The circuit described here overcomes the above-mentioned problem. It has a small memory which enables it to automatically switch ‘on’ and switch ‘off’ the lights in a desired fashion. The circuit uses two LDRs which are placed one after another (separated by a distance of say half a metre) so that they may separately sense a person going into the room or coming out of the room. Outputs of the two LDR sensors, after processing, are used in conjunction with a bicolour LED in such a fashion that when a person gets into the room it emits green light and when a person goes out of the room it emits red light, and vice versa.

These outputs are simultaneously applied to two counters. One of the counters will count as +1, +2, +3 etc when persons are getting into the room and the other will count as -1, -2, -3 etc when persons are getting out of the room. These counters make use of Johnson decade counter CD4017 ICs. The next stage comprises two logic ICs which can combine the outputs of the two counters and determine if there is any person still left in the room or not. Since in the circuit LDRs have been used, care should be taken to protect them from ambient light. If desired, one may use readily available IR sensor modules to replace the LDRs.

The sensors are installed in such a way that when a person enters or leaves the room, he intercepts the light falling on them sequentially—one after the other. When a person enters the room, first he would obstruct the light falling on LDR1, followed by that falling on LDR2. When a person leaves the room it will be the other way round. In the normal case light keeps falling on both the LDRs, and as such their resistance is low (about 5 kilo-ohms). As a result, pin 2 of both timers (IC1 and IC2), which have been configured as monostable flip-flops, are held near the supply voltage (+9V).

When the light falling on the LDRs is obstructed, their resistance becomes very high and pin 2 voltages drop to near ground potential, thereby triggering the flip-flops. Capacitors across pin 2 and ground have been added to avoid false triggering due to electrical noise. When a person enters the room, LDR1 is triggered first and it results in triggering of monostable IC1.

The short output pulse immediately charges up capacitor C5, forward biasing transistor pair T1-T2. But at this instant the collectors of transistors T1 and T2 are in high impedance state as IC2 pin 3 is at low potential and diode D4 is not conducting. But when the same person passes LDR2, IC2 monostable flip-flop is triggered. Its pin 3 goes high and this potential is coupled to transistor pair T1-T2 via diode D4. As a result transistor pair T1-T2 conducts because capacitor C5 retains the charge for some time as its discharge time is controlled by resistor R5 (and R7 to an extent).

Thus green LED portion of bi-colour LED is lit momentarily. The same output is also coupled to IC3 for which it acts as a clock. With entry of each person IC3 output (high state) keeps advancing. At this stage transistor pair T3-T4 cannot conduct because output pin 3 of IC1 is no longer positive as its output pulse duration is quite short and hence transistor collectors are in high impedance state. When persons leave the room, LDR2 is triggered first followed by LDR1.

Since the bottom half portion of circuit is identical to top half, this time with the departure of each person red portion of bi-colour LED is lit momentarily and output of IC4 advances in the same fashion as in case of IC3. The outputs of IC3 and those of IC4 (after inversion by inverter gates N1 through N4) are ANDed by AND gates (A1 through A4) are then wire ORed (using diodes D5 through D8). The net effect is that when persons are entering, the output of at least one of the AND gates is high, causing transistor T5 to conduct and energise relay RL1.

The bulb connected to the supply via N/O contact of relay RL1 also lights up. When persons are leaving the room, and till all the persons who entered the room have left, the wired OR output continues to remain high, i.e. the bulb continues to remains ‘on,’ until all persons who entered the room have left. The maximum number of persons that this circuit can handle is limited to four since on receipt of fifth clock pulse the counters are reset.

The capacity of the circuit can be easily extended for up to nine persons by removing the connection of pin 1 from reset pin (15) and utilising Q1 to Q9 outputs of CD4017 counters. Additional inverters, AND gates and diodes will, however, be required.

Running Message Display

2009-04-08T00:14:00.000-07:00

Light emitting diodes are advan- tageous due to their smaller size, low current consumption and catchy colours they emit. Here is a running message display circuit wherein the letters formed by LED arrangement light up progressively. Once all the letters of the message have been lit up, the circuit gets reset. The circuit is built around Johnson decade counter CD4017BC (IC2). One of the IC CD4017BE’s features is its provision of ten fully decoded outputs, making the IC ideal for use in a whole range of sequencing operations. In the circuit only one of the outputs remains high and the other outputs switch to high state successively on the arrival of each clock pulse.

The timer NE555 (IC1) is wired as a 1Hz astable multivibrator which clocks the IC2 for sequencing operations. On reset, output pin 3 goes high and drives transistor T7 to ‘on’ state. The output of transistor T7 is connected to letter ‘W’ of the LED word array (all LEDs of letter array are connected in parallel) and thus letter ‘W’ is illuminated. On arrival of first clock pulse, pin 3 goes low and pin 2 goes high. Transistor T6 conducts and letter ‘E’ lights up. The preceding letter ‘W’ also remains lighted because of forward biasing of transistor T7 via diode D21. In a similar fashion, on the arrival of each successive pulse, the other letters of the display are also illuminated and finally the complete word becomes visible. On the following clock pulse, pin 6 goes to logic 1 and resets the circuit, and the sequence repeats itself. The frequency of sequencing operations is controlled with the help of potmeter VR1.

The display can be fixed on a veroboard of suitable size and connected to ground of a common supply (of 6V to 9V) while the anodes of LEDs are to be connected to emitters of transistors T1 through T7 as shown in the circuit. The above circuit is very versatile and can be wired with a large number of LEDs to make an LED fashion jewellery of any design. With two circuits connected in a similar fashion, multiplexing of LEDs can be done to give a moving display effect.

Electronic Scoring Game

2009-04-08T00:06:00.000-07:00

You can play this game alone or with your friends. The circuit comprises a timer IC, two decade counters and a display driver along with a 7-segment display.

The game is simple. As stated above, it is a scoring game and the competitor who scores 100 points rapidly (in short steps) is the winner. For scoring, one has the option of pressing either switch S2 or S3. Switch S2, when pressed, makes the counter count in the forward direction, while switch S3 helps to count downwards. Before starting a fresh game, and for that matter even a fresh move, you must press switch S1 to reset the circuit. Thereafter, press any of the two switches, i.e. S2 or S3.

On pressing switch S2 or S3, the counter’s BCD outputs change very rapidly and when you release the switch, the last number remains latched at the output of IC2. The latched BCD number is input to BCD to 7-segment decoder/driver IC3 which drives a common-anode display DIS1. However, you can read this number only when you press switch S4. The sequence of operations for playing the game between, say two players ‘X’ and ‘Y’, is summarised below:

Player ‘X’ starts by momentary pressing of reset switch S1 followed by pressing and releasing of either switch S2 or S3. Thereafter he presses switch S4 to read the display (score) and notes down this number (say X1) manually.
Player ‘Y’ also starts by momentary pressing of switch S1 followed by pressing of switch S2 or S3 and then notes down his score (say Y1), after pressing switch S4, exactly in the same fashion as done by the first player.
Player ‘X’ again presses switch S1 and repeats the steps shown in step 1 above and notes down his new score (say, X2). He adds up this score to his previous score. The same procedure is repeated by player ‘Y’ in his turn.
The game carries on until the score attained by one of the two players totals up to or exceeds 100, to be declared as the winner.

Several players can participate in this game, with each getting a chance to score during his own turn. The assembly can be done using a multipurpose board. Fix the display (LEDs and 7-segment display) on top of the cabinet along with the three switches. The supply voltage for the circuit is 5V

Wiper Speed Control

2009-04-07T23:34:00.000-07:00

A continuously working wiper in a car may prove to be a nuisance, especially when it is not raining heavily. By using the circuit described here one can vary sweeping rate of the wiper from once a second to once in ten seconds. The circuit comprises two timer NE555 ICs, one CD4017 decade counter, one TIP32 driver transistor, a 2N3055 power transistor (or TIP3055) and a few other discrete components.

Timer IC1 is configured as a mono- stable multivibrator which produces a pulse when one presses switch S1 momentarily. This pulse acts as a clock pulse for the decade counter (IC2) which advances by one count on each successive clock pulse or the push of switch S1. Ten presets (VR1 through VR10), set for different values by trial and error, are used at the ten outputs of IC2.

But since only one output of IC2 is high at a time, only one preset (at selected output) effectively comes in series with timing resistors R4 and R5 connected in the circuit of timer IC3 which functions in astable mode. As presets VR1 through VR10 are set for different values, different time periods (or frequencies) for astable multivibrator IC3 can be selected.

The output of IC3 is applied to pnp driver transistor T1 (TIP32) for driving the final power transistor T2 (2N3055) which in turn drives the wiper motor at the selected sweep speed. The power supply for the wiper motor as well as the circuit is tapped from the vehicle’s battery itself. The duration of monostable multivibrator IC1 is set for a nearly one second period.

Ultrasonic Pest Repellant

2009-04-07T23:27:00.000-07:00

This electronic circuit is an ultrasonic pest repellant are repelled by variabble ultrasonic frequency in the range of 30 kHz to 50 kHz. Thus to increase the effectiveness, frequency of ultrasonic oscillator has to be continuously varied between certain limits.

By using this circuit design, frequency of emission of ultrasonic sound is continuously varied step-by-step automatically. Here five steps of variation are used but the same can be extended up to 10 steps, if desired. For each clock pulse output from op-amp IC1 CA3130 (which is wired here as a low-frequency square wave oscillator), the logic 1 output of IC2 CD4017 (which is a well-known decade counter) shifts from Q0 to Q4 (or Q0 to Q9).

Five presets VR2 through VR6 (one each connected at Q0 to Q4 output pins) are set for different values and connected to pin 7 of IC3 (NE555) electronically. VR1 is used to change clock pulse rate. IC3 is wired as an astable multivibrator operating at a frequency of nearly 80 kHz. Its output is not symmetrical. IC4 is CD4013, a D-type flip-flop which delivers symmetrical 40kHz signals at its Q and Q outputs which are amplified in push-pull mode by transistors T1, T2, T3 and T4 to drive a low-cost, high-frequency piezo tweeter.

For frequency adjustments, you may use an oscilloscope. It can be done by trial and error also if you do not have an oscilloscope. This pest repeller would prove to be much more effective than those published earlier because here ultrasonic frequency is automatically changed to cover different pests and the power output is also sufficiently high. If you want low-power output in 30-50 kHz ultrasonic frequency range then the crystal transducer may be directly connected across Q and Q outputs of IC4 (transistor amplifier is not necessary).

Power Supply Circuit 12-15 Volt 20A

2009-04-07T22:52:00.000-07:00

Output voltage of the power supply circuit is adjustable from fine potensiometer from 12V to 15v. It is suitable for all 12V power supply devices, or devices which are normally connected to a 12V battery or a vehicle with a 12V power supply system. This tension is usually 13.8 V.

For above reason, The Power Supply is also set to this tension, all right, however, any voltage from 12V to 14V. In this case, the tension is set somewhere around 13.6 V. To provide tension resistance in addition to voltage regulator 78S12. Instead potentiometer 100R inserted resistor 56R.

The scheme of the power supply is simple, but it is partly taken from some of the schemes taken up in the past. The material used is easily obtainable in electronic component shops, and this was the condition when I started to design this power supply.

ICL7107 Digital LED Voltmeter

2009-04-07T22:28:00.000-07:00

This circuit is a digital voltmeter with LED display. It's ideal to use for measuring the output voltage of your DC power supply. It includes a 3.5-digit LED display with a negative voltage indicator. It measures DC voltages from 0 to 199.9V with a resolution of 0.1V. The voltmeter is based on single ICL7107 chip and may be fitted on a small 3cm x 7cm printed circuit board. The circuit should be supplied with a 5V voltage supply and consumes only around 25mA.

The use of 7805 5V voltage regulator is highly recommended to prevent the damage of ICL7107, 555 ICs and to extend the operating voltages.

Parts list of The Digital LED Voltmeter:

R1 = 8K2 R1 = 8K2
R2 = 47K / 470K R2 = 47k / 470K
R3 = 100K R3 = 100K
R4 = 2K R4 = 2K
R5, R6 = 47K R5, R6 = 47k
R7 = 0R / 4K7 R7 = 0R / 4K7
R8 = 560R R8 = 560R
C1,C5, C6, C8, C9 = 100n C1, C5, C6, C8, C9 = 100n
C2 = 470n / 47n C2 = 470n / 47n
C3 = 220n C3 = 220n
C4 = 100p C4 = 100p
C7 = 10-22u C7 = 10-22U
D1, D2 = 1N4148 D1, D2 = 1N4148
IC1 = ICL7107 IC1 = ICL7107
IC2 = NE555 IC2 = NE555
OPTO = CA 10 pin FTA = CA 10 pin

The digital LED voltmeter can also be configured to measure different voltage ranges and display higher voltage resolution.

Battery Low Voltage Beeper

2009-04-07T22:15:00.000-07:00

This electronic circuit is an alarm circuit for low battery condition. It provides an audible and visual low voltage warning for 12V battery powered devices. When the battery voltage is above the set point (typically 11V), the circuit is idle. If the battery voltage should fall below the set point, the LED will light and the speaker will emit a periodic beeping sound to warn of the impending loss of power. The circuit was designed for monitoring solar systems, but it could also be useful for automotive and other 12V applications.

How it works

U2 provides a 5V regulated voltage reference. U1 is wired as a comparator, it compares the fixed 5V regulated voltage to the voltage on the wiper of VR1, that is proportional to the 12V supply. When the supply drops below the set point, the output of U1 goes low, turning on Q1 and powering the beeper and the LED.

The beeper consists of U4, a tone generator, and U3, a low duty cycle pulse generator. The tone can be changed by adjusting R7, the beep rate can be changed by adjusting R5. A small amount of hysteresis is provided by R1 and the current through LED1 and the beeper, this separates the on and off points for the circuit.

U2 provides a 5V regulated voltage reference. U1 is wired as a comparator, it compares the fixed 5V regulated voltage to the voltage on the wiper of VR1, that is proportional to the 12V supply. When the supply drops below the set point, the output of U1 goes low, turning on Q1 and powering the beeper and the LED.

The beeper consists of U4, a tone generator, and U3, a low duty cycle pulse generator. The tone can be changed by adjusting R7, the beep rate can be changed by adjusting R5. A small amount of hysteresis is provided by R1 and the current through LED1 and the beeper, this separates the on and off points for the circuit.

Use of Battery Low Voltage Beeper

Connect the circuit to the 12V source that you wish to monitor. Turn S1 on, if the battery voltage is above the set point, nothing should happen.

As the battery voltage drops below the set point, the LED will light and a periodic beeping will come from the speaker. If the beeping becomes annoying, turn off S1. Be sure to charge the battery soon, excessive discharging will shorten the life of most rechargeable batteries.

Car Anti-Theft Wireless Alarm

2009-04-07T22:11:00.000-07:00

This alarm circuit is an anti- theft wireless alarm can be used with any vehicle having 6- to 12-volt DC supply system. The mini VHF FM radio-controlled, FM transmitter is fitted in the vehicle at night when it is parked in the car porch or car park.

The receiver unit of the wireless alarm uses an CXA1019, a single IC-based FM radio module, which is freely available in the market at reasonable rate, is kept inside. Receiver is tuned to the transmitter's frequency. When the transmitter is on and the signals are being received by FM radio receiver, no hissing noise is available at the output of receiver. Thus transis- tor T2 (BC548) does not conduct. This results in the relay driver transistor T3 getting its forward base bias via 10k resistor R5 and the relay gets energised.

When an intruder tries to drive the car and takes it a few metres away from the car porch, the radio link betw- een the car (transmitter) and alarm (receiver) is broken. As a result FM radio module gene-rates hissing noise. Hissing AC signals are coupled to relay switching circ- uit via audio transformer. These AC signals are rectified and filtered by diode D1 and capacitor C8, and the resulting positive DC voltage provides a forward bias to transistor T2. Thus transistor T2 conducts, and it pulls the base of relay driver transistor T3 to ground level. The relay thus gets de-activated and the alarm connected via N/C contacts of relay is switched on.

If, by chance, the intruder finds out about the wireless alarm and disconnects the transmitter from battery, still remote alarm remains activated because in the absence of signal, the receiver continues to produce hissing noise at its output. So the burglar alarm is fool-proof and highly reliable. (Ed: You may have some problem catching the thief, though, if he decides to run away with your vehicle_in spite of the alarm!)

2-Way Electronic Crossover Network

2009-04-07T02:10:00.000-07:00

The electronic crossover featured here is an 18dB / octave unit, and has the crossover frequency centred on 300Hz. The frequency may be changed by increasing (or decreasing) resistor / capacitor values.

Increasing capacitance or resistance - Reduces frequency

Doubling the capacitance or resistance halves the frequency

Reducing capacitance or resistance - Increases frequency

Halving the capacitance or resistance doubles the frequency

The values of resistance and capacitance (indicated with a * in the circuit diagram) in the filter are critical, and close tolerance components are mandatory. If you cannot obtain close tolerance capacitors, use a capacitance meter to select values within 5% of the indicated value. Use only 1% metal film resistors throughout. The 1uF coupling caps are not critical, and standard tolerance is Ok.

If the crossover frequency is changed, it is critical that the ratios of capacitor and resistor values are not varied. For example, if you wanted to halve the frequency, the resistors would become 22k and 102k (100k is only just acceptable. If the ratios are changed, the filter damping is also changed, and the behaviour at the crossover point will be unpredictable (causing a dip or peak in the frequency response).

The values you change to alter the crossover frequency are indicated with a * in the circuit diagram

Do not change the 10k resistors - they set the damping of the filter and strange happenings will befall s/he who fiddles indiscriminately.

the NE5532 Dual op-amp is used. This circuit can be operated from the same power supply as the Audio Preamp, featured elsewhere on these pages. Other dual opamps may also be used, depending on your preference.
The input is buffered by U1a (the second channel can use the other half of the op-amp), and fed to the two filter networks. Each filter is a 3rd order section, and has a gain of 2. The output of each section is fed (via a 1uF polyester capacitor) to the level control and output buffer stage.
In use, the output of the preamplifier is fed to the input of the crossover network, and the outputs are fed to their respective amplifiers. For more information on bi-amping, refer to the article "Bi-Amplification - Not quite magic (but close)" on these pages.

Be careful when adjusting the level controls, since it is easy to create a mismatch in levels between the amplifiers. I suggest that the controls be mounted on the rear panel, with their shafts cut off really short, and a slot cut into the end with a hacksaw. Once the adjustment is made, it should not require further changes in use. Make sure that the power amplifier volume controls (if fitted) are turned fully up, and try to set the crossover controls so somewhere between midway and 75%. This ensures plenty of scope for getting the levels right, and will ensure that the preamp settings are not radically different from their "pre-biamp" days.

Classic PIC Programmer

2009-04-07T00:49:00.000-07:00

PICs are small microprocessors containing RAM, ROM, and some peripherals. Almost no other parts are required to make a complete “embedded system”. They are readily available and well supported by the manufacturer, third party developers, and most importantly, users. This has led to their immense popularity.

Assembly:

The PC board design is fairly straightforward and can be made by laser printing to special paper or a page from TIME magazine, then ironing the image onto copper-clad board, then etching with ferric chloride. There are a few jumper wires. The power source needs to be at least 15 volts. A 12 volt DC adapter usually produces about 17 volts, so that's a good choice. Two 9-volt batteries in series will work too. Solder directly to the PC board or use a connector that mates with your power source. Pay attention to the direction of the voltage regulators because the plastic regulators are backward from the tab type. Substitute Japanese or European generic equivalents for the transistors and diodes, but remember that the pinouts will be different. A right angle PC mount DB-25M connector is specified, but a conventional solder-cup DB-25M connector works, see the picture how I did it.

Operation:

The programmer connects to the parallel printer port of your computer and requires external power. If you want to program a PIC you'll have a hex file created by your assembler or created by someone else(see my propeller clock). You will also need to drive the programmer with some software. Here are programs that run under DOS and Windows. Linux software for Intel-based computers is available elsewhere. Macintoshes do not have parallel ports and can't use this programmer. Do not insert the PIC to be programmed until you have power applied and have run the software, and the programming LED is not lit. The DOS software requires command line switches for fuse settings(unless in the chip's hex file)and also the environmental variable "set ppsetup=3" to be typed before running the program. The Windows software requires the driver "dtait.drv" to be in the \windows\system directory and also the line "PINAPI=DTAIT.DRV" added to the system.ini file. Tell the software you have 7407 chip and PNP transistors. These details are explained in the text files included with the software.

The Files:

DOS software by David Tait "pic84v05.zip".
Windows software v1.03 by Silicon Studio "picser.zip".

Programming newer PICs:

The whole “F” series can be programmed. You need to use newer software, like this cool software. The PIC16F627 and PIC16F628 are 18 pin devices and fit right in the socket, but you must make a ground connection to pin 10 to prevent LVP programming, a new feature this programmer does not use. Some people suggest using a 10K resistor to ground, if you are doing in-circuit programming that probably makes sense. Programming the bigger PICs, including the PIC16F872 through 16F877 requires fitting the correct(28 or 40 pin) socket and wiring the pins to the corresponding function. Remember to ground the LVP pin on these, too. The bigger PICs also have extra power and ground pins. These must all be used.
You must select the port your computer is using(usually 0378) and the type of programmer (P16PRO) and the type of buffers the programmer uses (non-inverting). The software is beta, but I have tested it and can testify it working on the 16F84 and 16F628 I tried. It only programs locations used in the hex file, so it is very fast. If your program is 250 bytes, only 250 bytes get programmed, but when I used my PicstartPlus to verify the chips I tried, it would show a verify error unless I blanked the chip first, although the chip functioned fine. Leaving those unused areas in the previously programmed state shouldn’t be a problem.
Propic2 keeps the power to the chip on while idle. This can be useful for “burn and crash” in-circuit programming. You’ll see the LED is lit. I don’t like to insert or remove the PIC when power is present, so I pull the power cord before I insert or remove the PIC.
propic2 software, in case the above link is broken.

Simple AV (audio / video) Wireless Transmitter

2009-04-07T00:40:00.000-07:00

This circuit provides you with wireless audio and visual transmission to a TV. The TV acts as a receiver, eliminating the need to buy a separate monitor. You can also hook it up to a VCR or CCD Camera, and even set up a remote CCTV security system!

Circuit Analysis:

Q3, VC1, C13, C16 and L3 all make up a colpitts oscillator circuit that fluctuates form 220~250MHz. You can regulate the frequency to any value within this threshold by tuning VC1 or L3. C13 modulates the signal rate. When the capacitance increases, so does the modulation. R9 and C16 bias the local oscillation. If you lower R9's frequency to 680W the oscillator's output level will increase.
Q2 and L2 act as a frequency doubler. C7, along with FCZ7S3R5 (IF transformer), the Q4 transistor, C14, C19 and R12 all make up the mixer. This mixer takes both audio and visual signals together and "mix" them into one and passes through RF Amplifier Q1 to transmit the signal to the antenna.

Testing:

Turning the blue component's trimmer on VC1 varies the frequency. When we turn the trimmer, the television's channel has to be changed accordingly. It is easier to tune the A/V Sender if you have a spectrum analyzer to help you find the correct frequencies. If the frequency is tuned to 474 MHz then this would be the equivalent of your TV's channel 14 UHF band.
The IF transformer is used to synchronize the audio and video frequency's level radio. If the TV's image is too blurry then you can adjust the IFT to fine-tune the image.
SVR1 controls the video signal input ratio, while SVR2 controls the audio portion. You can tune these components according to your needs.

Bench Power Supply Unit

2009-04-07T00:35:00.000-07:00

Here is a rather novel Power Supply Unit (PSU) for the workbench which can deliver 0-15 volts and up to 0-3 amperes. I have not shown any mains fuses as this I will leave to your own devices. You should as a minimum have a fuse in the primary and secondary of T1.

The circuit is a little cramped, but this is because I will also be posting the circuit to QRP@WW on the packet radio system. There are two resistors in the circuit marked "*see text" and these set the maximum output current limiter to a suitable value. Omit the 0R2 resistor and the PSU will deliver 100mA maximum. In this event the output TR8 can be almost any medium power transistor such as the 2N3053, BFY51 etc. It must be mounted on a heatsink no matter what output you need.

The 6R8 resistor may be an ordinary 1/4 watt resistor and is used on all versions of the PSU. The additional 0R2 (0.2 ohms) sets the maximum current limiting to about 3 amperes. If you only want a 0-1 ampere PSU then use 0.68 ohms. T1 is a simple transformer suitably rated for the output power you want and having a 15 volt secondary. This should give over 21v of DC when rectified (measured at D1 cathode - ground). The anode of D7 should also have -21 volts with respect to ground.

If you have problems getting a 15 volt transformer then you can use a 12-volt transformer and add a few extra turns to the secondary. Most transformers have enough room for you to thread another 20 turns of wire through the former to put in series with the secondary winding. By sure to use the same thickness of wire as used for the factory made secondary.

D1, D2, D3, D4, D5, D6, D7 and D8 are all 1N4001 or better for up to 500mA but 1N5401 should be used if you want up to 3 amperes. D9 and D10 may be almost any small silicon diode such as 1N4148 etc. The 1K0 potentiometer in parallel with D9 is the current set pot. The 47K pot is the output voltage pot. With the values shown you should be able to get a range of 0v to a little over 15v.

HOW IT WORKS

TR4 and TR5 compare the ground (0v) with the wiper of the 47K pot which should also be 0v. If the voltages are not equal then TR5 controls the output darlington (TR6 and TR7) until the voltages become equal. If the 47K pot is set to maximum (top) then the output voltage of the PSU must be zero in order to get a balance. If the output voltage pot is set to the bottom then the voltage balance will only occur if there is +15 volts on the top of the 47K pot; the other end of the 15K being -5v.

In the event TR7 draws too much current then the voltage drop accross the 6R8 resistor will turn on TR1 which in turn turns on TR2. TR2 sinks the 0v reference to -20 volts. The TIP41 is incapable of delivering a negative output voltage so the output remains at 0v. The base of TR1 is "lifted" by up to 0.7v due to the current flowing through D10. This overcomes the TR1 base-emitter voltage drop needed to turn it on. You can adjust the 22K resistor a little so that the current control goes all the way down to 0mA and no further. 5K6 is about the absolute minimum resistance you should use.

The power supply itself is very simple and is nothing more than a full-wave bridge rectifier. The AC from the transformer is also capacitively coupled to yet another rectifier to provide the negative voltage required to allow the PSU to be controlled down to 0 volts. TR3 regulates this to stabilise the negative reference voltage.

POSSIBLE PROBLEMS

In the event of self-oscillation of the PSU then under no circumstances put any capacitance accross any signal in the control path. This will just make the problem worse. The cure for this would be to add capacitance between the base/emitter of the controlling transistors TR4 and TR5.

6V to 12V Converter

2009-04-06T21:30:00.000-07:00

This inverter circuit can provide up to 800mA of 12V power from a 6V supply. For example, you could run 12V car accessories in a 6V (British?) car. The circuit is simple, about 75% efficient and quite useful. By changing just a few components, you can also modify it for different voltages.

Parts

R1, R4 ------ 2 2.2K 1/4W Resistor
R2, R3 ------ 2 4.7K 1/4W Resistor
R5 ------ 1 1K 1/4W Resistor
R6 ------ 1 1.5K 1/4W Resistor
R7 ------ 1 33K 1/4W Resistor
R8 ------ 1 10K 1/4W Resistor
C1,C2 ------ 2 0.1uF Ceramic Disc Capacitor
C3 ------ 1 470uF 25V Electrolytic Capcitor
D1 ------ 1 1N914 Diode
D2 ------ 1 1N4004 Diode
D3 ------ 1 12V 400mW Zener Diode
Q1, Q2, Q4 --- 3 BC547 NPN Transistor
Q3 ------ 1 BD679 NPN Transistor
L1 ------ 1 See Notes
MISC ------ 1 Heatsink For Q3, Binding Posts (For Input/Output), Wire, Board

Notes

L1 is a custom inductor wound with about 80 turns of 0.5mm magnet wire around a toroidal core with a 40mm outside diameter.
Different values of D3 can be used to get different output voltages from about 0.6V to around 30V. Note that at higher voltages the circuit might not perform as well and may not produce as much current. You may also need to use a larger C3 for higher voltages and/or higher currents.
You can use a larger value for C3 to provide better filtering.
The circuit will require about 2A from the 6V supply to provide the full 800mA at 12V.

Video Stabilizer/MacroVision Removal

2009-04-06T21:20:00.000-07:00

Have you ever attempted to copy a commercially produced video only to end up with a distorted and jumpy image? If so, then you have run afoul of MacroVision. MacroVision is the most popular copy protection scheme used on the majority of content distributed on VHS cassettes. Like all copy protection, it does nothing to discourage the real pirates and only annoys the user who may wish to create a legal copy for backup and archival purposes. This circuit can eliminate MacroVision encoding in both NTSC and PAL recordings.

Parts

R1, R2, R3, R4, R5, R7 -------- 6 68K 1/4W Resistor
R6 ---------------------------- 2 75 Ohm 1/4W Resistor
R8, R9 ------------------------ 2 100 Ohm 1/4W Resistor
R10 --------------------------- 1 2.2K 1/4W Resistor
R11 ---------------------------- 1 1.5K 1/4W Resistor
R12, R17 ---------------------- 2 470K 1/4W Resistor
R13, R16 ---------------------- 2 33K 1/4W Resistor
R14 --------------------------- 1 6.8K 1/4W Resistor
R15 --------------------------- 1 22K 1/4W Resistor
R18 --------------------------- 1 47K 1/4W Resistor
C1 --------------------------- 1 15uF Electrolytic Capacitor
C2 --------------------------- 1 220uF Electrolytic Capacitor
C3 --------------------------- 1 220pF Ceramic Disc Capacitor
C4 --------------------------- 1 0.0022uF Ceramic Disc Capacitor
C5, C7, C8, C9, C10, C11, C12, C13 7 0.1uF Ceramic Disc Capacitor
C6 -------------------------- 1 0.47uF Ceramic Disc Capacitor
D1-D9, D10-D18, D19-D27, D28, D29, D30, D31, D32, D33, D34, D35 35 1N4148 Diode
Q1 --------------------------- 1 BC548 NPN Transistor
Q2 --------------------------- 1 BC558 PNP Transistor
U1 --------------------------- 1 4040 12-Bit Ripple Counter
U2, U3 ------------------------ 2 4027 Dual J-K Flip Flop
U4 ---------------------------- 1 4011 Quad Two Input NAND Gate
U5 --------------------------- 1 TL802 Dual Op Amp
U6 --------------------------- 1 4053 Triple Two Channel Multiplexer
U7 --------------------------- 1 4069 Hex Inverter
J1-J9, J10-J18, J19-J27 ------- 27 Jumper
J28, J29 --------------------- 2 RCA Jack
MISC ---------------------- 1 PC Board, Wire, Sockes For ICs

Notes

The circuit was submitted via email by Jari Ekstrom. Jari noted that the author is Antti Paarlahti and that the circuit originally came from the author's website. This website is no longer accessible. However, some searching revealed that the author of the circuit also wrote The MacroVision FAQ. The email address listed for the author in the FAQ is invalid and the FAQ was last updated in late 1996. This circuit reproduced here without permission.
Before use, the circuits's jumper need to be set. Take a look at the table below:

MacroVision Type
Upper Start Line
Upper End Line

Lower Start Line
PAL 0x05 0x0F 0x126
NTSC 0x06 0x0E 0xFB

To set the jumpers, first convert the line numbers to binary. You will end up with three binary digits, one for each set of line numbers. Bit 0 is least significant, bit 8 is most significant. Now simply open the jumpers at the 0 bits and close the jumpers at the 1 bits.
Connect your video source to "Composite Video In". As the label suggests, this circuit accempts composite video signals only. The corrected video signal is sent to the "Composite Video Out" jack.
With the two "Video" terminals disconnected the circuit passes video through without modifying it. Jumping the "Video" terminals enables the MacroVision removal.
As with all circuits involving CMOS (4000 series) ICs, you must tie the unused inputs of those ICs to ground. This is not shown on the schematic for clarity.
Supply voltage is 12V.

Water Detector With Sump/Bilge Pump Controller

2009-04-06T21:13:00.000-07:00

This moisture detector with pump controller is built around the special purpose LM1830 IC. The LM1830 is designed to detect moisture by passing an AC current through a set of probes. An internal comparator compares the resistance of the probes to an internal reference. When the resistance between the probes is low (indicating the presence of water or other conductive liquid) then the IC triggers it's output. An AC signal avoids electroplating and corrosion problems that are common when DC is used. To provide an adjustable delay hysteresis to avoid cycling the pump, a timer based around the 555 is used a range of about 5 seconds to two minutes. In this way the pump will stay active for a set amount of time even if the fluid level is below that of the probes.

Parts

R1 ------ 1 470 Ohm 1/4 W Resistor
R2, R4 ---- 1 10K 1/4 W Resistor
R3 ------- 1 6.8K 1/4 W Resistor
R5 ------- 1 1 Meg Linear Taper Pot
R6 ------- 1 51K 1/4 W Resistor
C1 ------- 1 470uF 35V Electrolytic Capacitor
C2 ------- 1 0.001uF Ceramic Disc Capacitor
C3 ------- 1 0.05 uF Ceramic Disc Capacitor
C4 ------- 1 20uF 35V Electrolytic Capacitor
C5 ------- 1 6.8uF 35V Electrolytic Capacitor
C6, C8 ---- 2 0.01uF Ceramic Disc Capacitor
C7 ------- 1 100uF 35V Electrolytic Capacitor
U1 ------- 1 LM1830 Fluid Detection IC
U2 ------- 1 555 Timer
Q1 ------- 1 2N5305 NPN Power Transistor
D1, D2 ---- 2 1N4148 Signal Diode
D3 ------ 1 1N4002 Rectifier Diode
K1 ------- 1 Relay With 12V Coil (See Notes)
S1 ------- 1 SPST Switch
PROBES ---- 1 Stainless Steel Probes (See Notes)
MISC ----- 1 Case, Knob, Board, Wire, Sockets For U1 and U2

Notes

The probes should be made of stainless steel and mounted in a non-conductive fixture. Stainless steel nails, bolts or welding rod will make ideal probes. Good non-conductive materials are Lexan, plexiglass or plastic. The probes are to be placed at the liquid level at which you wish the pump to turn on.
K1 should be chosen according to the requirements of the pump. If you are running a 12V pump then common automotive relays will work fine. If you are running a mains powered pump then you will need to make sure the contacts are rated accordingly.
S1 provides a manual pump switch.
R5 adjusts the on-time of the pump once moisture has been detected. It is adjustable from approximately 5 seconds to approximately 2 minutes.
The probes should be cleaned periodically to assure reliable operation.

Stepper Motor Controller

2009-04-06T21:07:00.000-07:00

The circuit is very simple and inexpensive. This is good thing because most commercial stepper motor controller ICs are quite expensive. This circuit is built from standard components and can easily be adapted to be controlled by a computer. If you use cheap surplus transistors and stepper motor, the price of the circuit can be kept to under $10.

Parts

R1, R2 ,R3, R4 ------ 4 1K 1/4W Resistor
D1, D2, D3, D4 ------ 4 1N4002 Silicon Diode
Q1, Q2, Q3, Q4 ------ 4 TIP31 NPN Transistor (See Notes) TIP41, 2N3055
U1 ----------------- 1 4070 CMOS XOR Integrated Circuit
U2 ----------------- 1 4027 CMOS Flip-Flop
S1 ----------------- 1 SPDT Switch
MISC ----------------- 1 Case, Board, Wire, Stepper Motor

Notes

You should be able to substitute any standard (2N3055, etc.) power transistor for Q1-Q4.
Every time the STEP line is pulsed, the motor moves one step.
S1 changes the motors direction.

Pulse Width Modulation DC Motor Control

2009-04-06T21:03:00.000-07:00

Often, people attempt to control DC motors with a variable resistor or variable resistor connected to a transistor. While the latter approach works well, it generates heat and hence wastes power. This simple pulse width modulation DC motor control eliminates these problems. It controls the motor speed by driving the motor with short pulses. These pulses vary in duration to change the speed of the motor. The longer the pulses, the faster the motor turns, and vice versa.

Parts

R1 1 1 Meg 1/4W Resistor
R2 1 100K Pot
C1 1 0.1uF 25V Ceramic Disc Capacitor
C2 1 0.01uF 25V Ceramic Disc Capacitor
Q1 1 IRF511 MOSFET IRF620
U1 1 4011 CMOS NAND Gate
S1 1 DPDT Switch
M1 1 Motor (See Notes)
MISC 1 Case, Board, Heatsink, Knob For R2, Socket For U1

Notes

R2 adjusts the speed of the oscillator and therefore the speed of M1.
M1 can be any DC motor that operates from 6V and does not draw more than the maximum current of Q1. The voltage can be increased by connecting the higher voltage to the switch instead of the 6V that powers the oscillator. Be sure not to exceed the power rating of Q1 if you do this.
Q1 will need a heatsink.
Q1 in the parts list can handle a maximum of 5A. Use the IRF620 for 6A, if you need any higher.
This circuit is not a true pulse width modulation control. Because only the frequency of pulses varies, it is really pulse frequency modulation. This works, though not as well as true PWM.

Digital Keypad Combination Lock

2009-04-06T20:57:00.000-07:00

This simple circuit is the electronic version of the combination lock. Using the special purpose LS7220 digital lock IC, the circuit allows a 4 digit combination of your choice to activate a relay for a set period of time. This relay can then be used to trigger a lock solenoid, enable a starter button, open a motorized door, or many other tasks that require a momentary signal.

Parts

C1 ------- 1 1uF 25V Electrolytic Capacitor
C2 ------- 1 220uF 25V Electrolytic Capacitor
R1 ------- 1 2.2K 1/4W Resistor
Q1 ------- 1 2N3904 NPN Transistor 2N2222 (Substitute)
D1 ------- 1 1N4148 Rectifier Diode 1N4001-1N4007 (Substitute)
K1 ------- 1 12V SPDT Relay Any appropriate relay with 12V coil
U1 ------- 1 LS7220 Digital Lock IC
S1-S12 ---- 12 SPST Momentary Pushbutton Keypad (see notes)
HD1 ----- 1 12 Position Header

Notes

To set the combination, wire the appropriate switches to U1 pins 3, 4, 5 and 6 using the header. For example if S1 was connected to pin 3, S2 to pin 4, S3 to pin 5 and S4 to pin 6, the combination would be 1,2,3,4. Now wire all other unused switches across the header to pin 2 of U1. In this way you can create any 4 digit combination you want. Pin 2 is the reset pin, so connecting all unused keys to it assures that the entire combination must be reentered if an incorrect key is pressed.
When the appropriate combination is entered, the relay is activated for a period of time determined by C1. The 1uF capacitor specified in the parts list will result in an on-time of roughly 5 seconds. Increase the value of C1 to increase this time.
An easy way to make a keypad is to buy 12 PC board mount pushbuttons and then etch a PC board so that the buttons are in 4 rows of 3, similar to a telephone keypad. Place this in a case and then use a label maker or transfer letters to add your numbers to the tops of the pushbuttons. You can also use a pre made keypad but keep in mind that you need a pad which provides an output for each key. Most pads available have the keys connected to provide a row and column signal when they are pressed.

Air Flow Detector

2009-04-06T20:53:00.000-07:00

This simple circuit uses an incandescent lamp to detect airflow. With the filament exposed to air, a constant current source is used to slightly heat the filament. As it is heated, the resistance increases. As air flows over the filament it cools down, thus lowering it's resistance. A comparator is used to detect this difference and light an LED. With a few changes, the circuit can be connected to a meter or ADC to provide an estimation on the amount of air flow.

Parts

R1 ---- 1 100 Ohm 1/4W Resistor
R2 ---- 1 470 Ohm 1/4W Resistor
R3 ---- 1 10k 1/4W Resistor
R4 ---- 1 100K 1/4W Resistor
R5 ---- 1 1K 1/4W Resistor
C1 ---- 1 47uF Electrolytic Capacitor
U1 ---- 1 78L05 Voltage Regulator
U2 ---- 1 LM339 Op Amp
L1 ---- 1 #47 Incandescent lamp with glass removed (See "Notes")
D1 ---- 1 LED
MISC --- 1 Board, Wire, Sockets for ICs, etc.

Notes

The glass will have to be removed from L1 without breaking the filament. Wrap the glass in masking tape and it in a vise. Slowly crank down until the glass breaks, then remove the bulb and carefully peel back the tape. If the filament has broken, you will need another lamp.

AC Motor Speed Controller

2009-04-06T20:51:00.000-07:00

This AC motor speed controller can handle most universal type (brushed) AC motors and other loads up to about 250W. It works in much the same was a light dimmer circuit; by chopping part of the AC waveform off to effectively control voltage. Because of this functionality, the circuit will work for a wide variety of loads including incandescent light bulbs, heating elements, brushed AC motors and some transformers. The circuit tries to maintain a constant motor speed regardless of load so it is also ideal for power tools. Note that the circuit can only control brushed AC motors. Inductive motors require a variable frequency control.

Parts

R1 ------- 1 27K 1W Resistor
R2 ------- 1 10K 1/4W Resistor
R3 ------- 1 100K 1/4W Resistor
R4 ------- 1 33K 1/4W Resistor
R5 ------- 1 2.2K 1/4W Resistor
R6 ------- 1 1K 1/4W Resistor
R7 ------- 1 60K Ohm 1/4W Resistor
R8 ------- 1 3K Linear Taper Trim Pot
R9 ------- 1 5K Linear Taper Pot
R10 ------- 1 4.7K Linear Taper Trim Pot
R11 ------- 1 3.3K 1/4W Resistor
R12 ------- 1 100 Ohm 1/4W Resistor
R13 ------- 1 47 Ohm 1W Resistor (See Notes)
C1, C3 ---- 2 0.1uF Ceramic Disc Capacitor
C2 ------- 1 100uF 50V Electrolytic Capacitor
D1 ------- 1 6V Zener Diode
Q1 ------- 1 2N2222 NPN Transistor 2N3904
SCR1 ----- 1 ECG5400
TR1 ----- 1 TRIAC (See Notes)
U1 -------- 1 DIAC Opto-Isolator (See Notes)
BR1, BR2 -- 2 5A 50V Bridge Rectifier
T1 ------- 1 Transformer (See Notes)
MISC ----- 1 PC Board, Case, Line Cord, Socket For U1, Heatsinks

Notes

TR1 must be chosen to match the requirements of the load. Most generic TRIACs with ratings to support your load will work fine in this circuit. If you find a TRIAC that works well, feel free to leave a comment.
U1 must be chosen to match the ratings of TR1. Most generic DIAC based opto-isolators will work fine. If you have success with a specific part, feel free to leave a comment.
T1 is any small transformer with a 1:10 turns ratio. The circuit is designed to run on 120V so a 120V to 12V transformer will work. Alternately, you can wind T1 on a transformer core using a primary of 25 turns, a secondary of 200 turns, and 26 gauge magnet wire.
R9 is used to adjust motor speed. R10 is a trim pot used to fine tune the governing action of the circuit. R8 fine tunes the feedback circuit to adjust for proper voltage at the gate of SCR1. It should be adjusted to just past the minimum point at which the circuit begins to operate.
R13 must be chosen to match the load. Generally, larger loads will require a smaller value.
Since this circuit is not isolated from mains, it must be built in an insulated case.

Colour (Sound) Organ

2009-04-06T20:36:00.000-07:00

Anyone who has been to a night club, concert or school dance has probobly seen a colour organ. Colour organs cause lights to blink and flash to music from your TV, stereo, guitar and even your own voice. The colour organ presented here needs no connection to the sound source, it picks up sound from its built in microphone.

Schematics

PCB Layout

Parts

C1 --------- 1 22uf 250V Electrolytic Capacitor
C2 --------- 1 22uf 250V Electrolytic Capacitor
C3 --------- 1 0.1uf Disc Capacitor
C4 --------- 1 0.01uf Disc Capacitor
C5 --------- 1 0.0047uf Disc Capacitor
R1 --------- 1 47K 1/2 W Resistor
R2, R4 ----- 2 6.8K 1/2 W Resistor
R3, R5 ----- 2 1M 1/2 W Resistor
R6 ----- 1 3.3K 1/2 W Resistor
R7, R8, R9 -- 3 1K 1/2 W Resistor
R10, R11, R12 3 10K Pot
D1 --------- 1 1N4004 Diode
Q1, Q2 ----- 2 2N3904 NPN Transistor 2N2222
Q3, Q4, Q5 - 3 106B1 SCR Teccor S2003LS1
T1 --------- 1 10K:600 Ohm Audio Transformer
S1 --------- 1 SPDT Switch
J1, J2, J3 -- 3 AC Socket
MISC ------- 1 AC Line Cord, Crystal Microphone, Case, Wire

Notes

R10, R11 and R12 control the response of the different lights.
The circuit can handle up to 300 watts per channel.
This circuit is NOT isolated from the 115 Volt line. If it is used with the case opened or not installed in a case, you could recieve a bad shock or be killed.
You can also use the Teccor S2003LS1 SCR for SCR1. These give better sensitivity and brightness than the 106B1 units.

3 Channel Spectrum Analyzer

2009-04-06T20:29:00.000-07:00

This 3 channel 15 LED spectrum analyzer is the perfect addition to any audio amp project. It produces fantastic displays on three LED bars that can be individually adjusted for any particular frequency range. The circuit will take line level output from most any audio source, and operates on 12V DC. This means that it can even be run in a car.

Part

R1 ------------------------ 1 100K 1/4W Resistor
R2 ------------------------ 1 820K 1/4W Resistor
R3, R14, R16, R18 ---------- 4 2.2 Meg 1/4W Resistor
R4, R5, R6 ---------------- 3 22K Pot
R7, R8, R9, R25, R27, R29 -- 6 10K 1/4W Resistor
R10, R11, R12 -------------- 3 680 Ohm 1/4W Resistor
R13. R15, R17 -------------- 3 580K 1/4W Resistor
R19, R20, R21 -------------- 3 39K 1/4W Resistor
R22, R23, R24 -------------- 3 47K 1/4W Resistor
R26, R28, R30 -------------- 3 33 Ohm 1/4W Resistor
C1, C5, C6 -------------- 3 0.012uF Polystyrene Capacitor
C2, C9, C10, C11 ----------- 4 3.3uF Electrolytic Capacitor
C3, C4 -------------------- 2 0.0022uF Polystyrene Capacitor
C7, C8 -------------------- 2 47nF Polystyrene Capacitor
C12, C13, C14 -------------- 3 0.47uF Electrolytic Capacitor
C15, C16, C17 -------------- 3 22uF Electrolytic Capacitor
D1, D2, D3 ----------------- 3 1N4002 Silicon Diode
D4, D5, D6, D8, D8 --------- 5 Green LED
D10, D11, D12, D13, D14 ---- 5 Amber LED
D16, D17, D18, D19, D20 ---- 5 Red LED
U1 ------------------------ 1 LM3900 Quad Op Amp
U2, U3, U4 ----------------- 3 AN6884 Bar Graph IC
MISC -------------------- 1 Board, Wires, Sockets For ICs

Notes:

The circuit expects line level inputs. If you connect it to speaker level, you will have to readjust the circuit every time you change the volume.
After the circuit is connected, apply power and signal. Now adjust the pots until the corresponding group of LEDs reacts.

12VDC Fluorescent Lamp Driver

2009-04-06T18:43:00.000-07:00

A number of people have been unable to find the transformer needed for the Black Light project, so I looked around to see if I could find a fluorescent lamp driver that does not require any special components. Here it is. It uses a normal 120 to 6V stepdown transformer in reverse to step 12V to about 350V to drive a lamp without the need to warm the filaments.

Parts

C1 ----- 1 100uf 25V Electrolytic Capacitor

C2,C3 -- 2 0.01uf 25V Ceramic Disc Capacitor
C4 ----- 1 0.01uf 1KV Ceramic Disc Capacitor
R1 ----- 1 1K 1/4W Resistor
R2 ----- 1 2.7K 1/4W Resistor
Q1 ----- 1 IRF510 MOSFET
U1 ----- 1 TLC555 Timer IC
T1 ----- 1 6V 300mA Transformer
LAMP --- 1 4W Fluorescent Lamp
MISC --- 1 Board, Wire, Heatsink For Q1

Notes

Q1 must be installed on a heat sink.
A 240V to 10V transformer will work better then the one in the parts list. The problem is that they are hard to find.
This circuit can give a nasty (but not too dangerous) shock. Be careful around the output leads.

Computer Controlled Frequency Counter/Logic Probe

2009-04-06T18:35:00.000-07:00

This circuit is a stable frequency counter accurate to 5 significant digits. The range is 0 - 30MHz with an input sensitivity of greater then 100mV. The probe connects to the PC serial port. So by using the crystal oscillator already present on your PC serial card and software calibration, the Probes' external circuitry is kept to a minimum. Probe 9 can also be used as a logic probe/analyzer using included software (LPROBE92.EXE).

Parts

R1,R2,R3,R4---- 4 100K 1/4W Resistor
R5 ------------ 1 10M 1/4W Resistor
R6, R7 ------- 2 3.3K 1/4W Resistor
R8 ----------- 1 390 Ohm 1/4W Resistor
R9 ----------- 1 1M 1/4W Resistor
C1, C4 ------- 2 0.1uF Ceramic Disc Capacitor
C2, C3 ------- 2 100uF 16V Electrolytic Capacitor
D1 ----------- 1 1N4148 Signal Diode Any 200mA silicon signal diode
D2, D3 ------- 2 3.3V Zener Diode
D4 ----------- 1 6.2V Zener Diode
U1 ----------- 1 74HC00 Quad Highspeed NAND Gate
U2, U3, U4 ---- 3 4021 8 Stage Shift Register
U5 ----------- 1 74HC393 Dual Highspeed 4 Bit Counter
U6 ----------- 1 4040 12 Stage Binary Counter
MISC --------- 1 PC Board, Wire, Suitable Probe, DB9/DB25 Connector

The software to use this probe can be downloaded using the following link. Note that this software is compiled for Intel x86 platforms and runs under DOS, Win95, Win98 and WinMe. It does not run under any Windows version based on NT including Windows NT 3.51, WinNT4, Win2K, Win2K3, WinXP and Windows Vista. This is because NT based operating systems do not allow direct hardware access.

Probe Software, Zipped, 19K

SETPROBE.EXE is the frequency counter calibration program. To give accurate readings the Probe must be calibrated to your PC. SETPROBE.EXE calculates the constant error correction factor for the particular PC serial card the probe is to be used on. The frequency counter corrects for this slight constant error in crystal frequency by using the correction factor contained in PROBE.DAT. To calculate this correction factor, a reliable oscillator of known frequency (eg 2MHz Crystal Oscillator) is required. When CALIBRAT.EXE is run, the Probe will sample the frequency and then ask for the true frequency value in HZ. The frequency entered must be to 1 Hz accuracy (no decimal points) or an error will occur (for example "200123" not "200123.34" or "2003.421 kHz"). The program then calculates constant error correction factor and stores it to PROBE.DAT. Calibration is only necessary once.
LPROBE92.EXE is the logic analysis program . Logic states are displayed in real time. This program runs best under DOS (not a DOS window). The sampling speed is adjusted by using the left and right arrow keys.

The three triggering modes are:

TRIG: Starts each scan (left-right of screen) on a negative going edge of logic signal.
KEY TRIG: Waits for a key to be pressed before beginning each scan.
FREE RUNNING: Not triggered.

To Toggle between these use the UP / DOWN arrow keys. To quit from LPROBE press escape.
FPROBE92.EXE is the frequency counter program. The measured frequency is displayed in Hz with commas indicating KHz and MHz. To quit from FPROBE press any key.
Serial port pinouts are as follows.

Park-Aid

2009-04-02T01:26:00.000-07:00

Three LEDs signal bumper-barrier distance
Infra-red operation, indoor use

This circuit was designed as an aid in parking the car near the garage wall when backing up. LED D7 illuminates when bumper-wall distance is about 20 cm., D7+D6 illuminate at about 10 cm. and D7+D6+D5 at about 6 cm. In this manner you are alerted when approaching too close to the wall.

All distances mentioned before can vary, depending on infra-red transmitting and receiving LEDs used and are mostly affected by the color of the reflecting surface. Black surfaces lower greatly the device sensitivity. Obviously, you can use this circuit in other applications like liquids level detection, proximity devices etc.

Parts:

R1_____________10K 1/4W Resistor
R2,R5,R6,R9_____1K 1/4W Resistors
R3_____________33R 1/4W Resistor
R4,R11__________1M 1/4W Resistors
R7______________4K7 1/4W Resistor
R8______________1K5 1/4W Resistor
R10,R12-R14_____1K 1/4W Resistors
C1,C4___________1µF 63V Electrolytic or Polyester Capacitors
C2_____________47pF 63V Ceramic Capacitor
C3,C5_________100µF 25V Electrolytic Capacitors
D1_____________Infra-red LED
D2_____________Infra-red Photo Diode (see Notes)
D3,D4________1N4148 75V 150mA Diodes
D5-7___________LEDs (Any color and size)
IC1_____________555 Timer IC
IC2___________LM324 Low Power Quad Op-amp
IC3____________7812 12V 1A Positive voltage regulator IC

Circuit operation:

IC1 forms an oscillator driving the infra-red LED by means of 0.8mSec. pulses at 120Hz frequency and about 300mA peak current. D1 & D2 are placed facing the car on the same line, a couple of centimeters apart, on a short breadboard strip fastened to the wall. D2 picks-up the infra-red beam generated by D1 and reflected by the surface placed in front of it. The signal is amplified by IC2A and peak detected by D4 & C4. Diode D3, with R5 & R6, compensates for the forward diode drop of D4. A DC voltage proportional to the distance of the reflecting object and D1 & D2 feeds the inverting inputs of three voltage comparators. These comparators switch on and off the LEDs, referring to voltages at their non-inverting inputs set by the voltage divider resistor chain R7-R10.

Circuit modification:

A circuit modification featuring an audible alert instead of the visual one is available here: Park-Aid Modification

Notes:

Power supply must be regulated (hence the use of IC3) for precise reference voltages. The circuit can be fed by a commercial wall plug-in adapter, having a DC output voltage in the range 12-24V.
Current drawing: LEDs off 40mA; all LEDs on 60mA @ 12V DC supply.
The infra-red Photo Diode D2, should be of the type incorporating an optical sunlight filter: these components appear in black plastic cases. Some of them resemble TO92 transistors: in this case, please note that the sensitive surface is the curved, not the flat one.
Avoid sun or artificial light hitting directly D1 & D2.
If your car has black bumpers, you can line-up the infra-red diodes with the (mostly white) license or number plate.
It is wiser to place all the circuitry near the infra-red LEDs in a small box. The 3 signaling LEDs can be placed far from the main box at an height making them well visible by the car driver.
The best setup is obtained bringing D2 nearer to D1 (without a reflecting object) until D5 illuminates; then moving it a bit until D5 is clearly off. Usually D1-D2 optimum distance lies in the range 1.5-3 cm.
If you are needing a simpler circuit of this kind driving a LED or a relay, click Infra-red Level Detector

Speed-limit Alert

2009-04-02T01:19:00.000-07:00

Wireless portable unit
Adaptable with most internal combustion engine vehicles

This circuit has been designed to alert the vehicle driver that he/she has reached the maximum fixed speed limit (i.e. in a motorway). It eliminates the necessity of looking at the tachometer and to be distracted from driving. There is a strict relation between engine's RPM and vehicle speed, so this device controls RPM, starting to beep and flashing a LED once per second, when maximum fixed speed is reached. Its outstanding feature lies in the fact that no connection is required from circuit to engine.

Circuit operation:

IC1 forms a differential amplifier for the electromagnetic pulses generated by the engine sparking-plugs, picked-up by sensor coil L1. IC2A further amplifies the pulses and IC2B to IC2F inverters provide clean pulse squaring. The monostable multivibrator IC3A is used as a frequency discriminator, its pin 6 going firmly high when speed limit (settled by R11) is reached. IC3B, the transistors and associate components provide timings for the signaling part, formed by LED D5 and piezo sounder BZ1. D3 introduces a small amount of hysteresis.

Parts:

R1,R2,R19_______1K 1/4W Resistors
R3-R6,R13,R17_100K 1/4W Resistors
R7,R15__________1M 1/4W Resistors
R8_____________50K 1/2W Trimmer Cermet
R9____________470R 1/4W Resistor
R10___________470K 1/4W Resistor
R11___________100K 1/2W Trimmer Cermet (see notes)
R12___________220K 1/4W Resistor (see notes)
R14,R16________68K 1/4W Resistors
R18____________22K 1/4W Resistor
R20___________150R 1/4W Resistor (see notes)
C1,C7_________100µF 25V Electrolytic Capacitors
C2,C3_________330nF 63V Polyester Capacitors
C4-C6___________4µ7 25V Electrolytic Capacitors
D1,D5______Red LEDs 3 or 5mm.
D2,D3________1N4148 75V 150mA Diodes
D4________BZX79C7V5 7.5V 500mW Zener Diode
IC1__________CA3140 or TL061 Op-amp IC
IC2____________4069 Hex Inverter IC
IC3____________4098 or 4528 Dual Monostable Multivibrator IC
Q1,Q2_________BC238 25V 100mA NPN Transistors
L1_____________10mH miniature Inductor (see notes)
BZ1___________Piezo sounder (incorporating 3KHz oscillator)
SW1____________SPST Slider Switch
B1_______________9V PP3 Battery (see notes)
Clip for PP3 Battery

Notes:

D1 is necessary at set-up to monitor the sparking-plugs emission, thus allowing to find easily the best placement for the device on the dashboard or close to it. After the setting is done, D1 & R9 can be omitted or switched-off, with battery savings.
During the preceding operation R8 must be adjusted for better results. The best setting of this trimmer is usually obtained when its value lies between 10 and 20K.
You must do this first setting when the engine is on but the vehicle is stationary.
The final simplest setting can be made with the help of a second person. Drive the vehicle and reach the speed needed. The helper must adjust the trimmer R11 until the device operates the beeper and D5. Reducing vehicle's speed the beep must stop.
L1 can be a 10mH small inductor usually sold in the form of a tiny rectangular plastic box. If you need an higher sensitivity you can build a special coil, winding 130 to 150 turns of 0.2 mm. enameled wire on a 5 cm. diameter former (e.g. a can). Extract the coil from the former and tape it with insulating tape making thus a stand-alone coil.
Current drawing is about 10mA. If you intend to use the car 12V battery, you can connect the device to the lighter socket. In this case R20 must be 330R.
Depending on the engine's cylinders number, R11 can be unable to set the device properly. In some cases you must use R11=200K and R12=100K or less.
If you need to set-up the device on the bench, a sine or square wave variable generator is required.
To calculate the frequency relation to RPM in a four strokes engine you can use the following formula:

Hz= (Number of cylinders * RPM) / 120.
For a two strokes engine the formula is: Hz= (Number of cylinders * RPM) / 60.
Thus, for a car with a four strokes engine and four cylinders the resulting frequency @ 3000 RPM is 100Hz.
Temporarily disconnect C2 from IC1 pin 6. Connect the generator output across C2 and Ground. Set the generator frequency to e.g. 100Hz and trim R11 until you will hear the beeps and LED D5 will start flashing. Reducing the frequency to 99 or 98 Hz, beeping and flashing must stop.
Please note that this circuit is not suited to Diesel engines.

60W MosFet Audio Amplifier

2009-04-02T01:09:00.000-07:00

High Quality, powerful unit: 90W into 4 Ohm load
Also suited as guitar or bass amplifier

To celebrate the hundredth design posted to this website, and to fulfil the requests of many correspondents wanting an amplifier more powerful than the 25W MosFet, a 60 - 90W High Quality power amplifier design is presented here. Circuit topology is about the same of the above mentioned amplifier, but the extremely rugged IRFP240 and IRFP9240 MosFet devices are used as the output pair, and well renowned high voltage Motorola's transistors are employed in the preceding stages.

The supply rails voltage was kept prudentially at the rather low value of + and - 40V. For those wishing to experiment, the supply rails voltage could be raised to + and - 50V maximum, allowing the amplifier to approach the 100W into 8 Ohm target: enjoy! A matching, discrete components, Modular Preamplifier design is available here: Modular Audio Preamplifier.

Parts:

R1______________47K 1/4W Resistor
R2_______________4K7 1/4W Resistor
R3______________22K 1/4W Resistor
R4_______________1K 1/4W Resistor
R5,R12,R13_____330R 1/4W Resistors
R6_______________1K5 1/4W Resistor
R7______________15K 1/4W Resistor
R8______________33K 1/4W Resistor
R9_____________150K 1/4W Resistor
R10____________500R 1/2W Trimmer Cermet
R11_____________39R 1/4W Resistor
R14,R15___________R33 2.5W Resistors
R16_____________10R 2.5W Resistor
R17_______________R22 5W Resistor (wirewound)
C1_____________470nF 63V Polyester Capacitor
C2_____________470pF 63V Polystyrene or ceramic Capacitor
C3______________47µF 63V Electrolytic Capacitor
C4,C8,C9,C11___100nF 63V Polyester Capacitors
C5______________10pF 63V Polystyrene or ceramic Capacitor
C6_______________1µF 63V Polyester Capacitor
C7,C10_________100µF 63V Electrolytic Capacitors
D1___________1N4002 100V 1A Diode
D2_____________5mm. Red LED
Q1,Q2,Q4_____MPSA43 200V 500mA NPN Transistors
Q3,Q5________BC546 65V 100mA NPN Transistors
Q6___________MJE340 200V 500mA NPN Transistor
Q7___________MJE350 200V 500mA PNP Transistor
Q8___________IRFP240 200V 20A N-Channel Hexfet Transistor
Q9___________IRFP9240 200V 12A P-Channel Hexfet Transistor

Power supply circuit diagram:

Parts:

R1_______________3K9 1W Resistor
C1,C2_________4700µF 63V Electrolytic Capacitors (See Notes)
C3,C4__________100nF 63V Polyester Capacitors
D1_____________400V 8A Diode bridge
D2_____________5mm. Red LED
F1,F2__________4A Fuses with sockets
T1_____________230V or 115V Primary, 30+30V Secondary 160VA Mains transformer
PL1____________Male Mains plug
SW1____________SPST Mains switch

Notes:

In the original circuit, a three-diode string was wired in series to R10. Two of these diodes are now replaced by a red LED in order to achieve improved quiescent current stability over a larger temperature range. Thanks to David Edwards of LedeAudio for this suggestion.
A small, U-shaped heatsink must be fitted to Q6 & Q7.
Q8 & Q9 must be mounted on large heatsinks.
Quiescent current can be measured by means of an Avo-meter wired in series to the positive supply rail and no input signal.
Set the Trimmer R10 to its minimum resistance.
Power-on the amplifier and adjust R10 to read a current drawing of about 120 - 130mA.
Wait about 15 minutes, watch if the current is varying and readjust if necessary.
The value suggested for C1 and C2 in the Power Supply Parts List is the minimum required for a mono amplifier. For optimum performance and in stereo configurations, this value should be increased: 10000µF is a good compromise.
A correct grounding is very important to eliminate hum and ground loops. Connect to the same point the ground sides of R1, R3, C2, C3 and C4 and the ground input wire. Connect R7 and C7 to C11 to output ground. Then connect separately the input and output grounds to the power supply ground.

Technical data:

Output power:

60 Watt RMS @ 8 Ohm (1KHz sinewave) - 90W RMS @ 4 Ohm

Sensitivity:

1V RMS input for 58W output

Frequency response:

30Hz to 20KHz -1dB

Total harmonic distortion @ 1KHz:

1W 0.003% 10W 0.006% 20W 0.01% 40W 0.013% 60W 0.018%

Total harmonic distortion @10KHz:

1W 0.005% 10W 0.02% 20W 0.03% 40W 0.06% 60W 0.09%
Unconditionally stable on capacitive loads

10W Audio Amplifier with Bass-boost

2009-04-02T01:05:00.000-07:00

High Quality, very simple design
No preamplifier required

This design is based on the 18 Watt Audio Amplifier, and was developed mainly to satisfy the requests of correspondents unable to locate the TLE2141C chip. It uses the widespread NE5532 Dual IC but, obviously, its power output will be comprised in the 9.5 - 11.5W range, as the supply rails cannot exceed ±18V.

As amplifiers of this kind are frequently used to drive small loudspeaker cabinets, the bass frequency range is rather sacrificed. Therefore a bass-boost control was inserted in the feedback loop of the amplifier, in order to overcome this problem without quality losses. The bass lift curve can reach a maximum of +16.4dB @ 50Hz. In any case, even when the bass control is rotated fully counterclockwise, the amplifier frequency response shows a gentle raising curve: +0.8dB @ 400Hz, +4.7dB @ 100Hz and +6dB @ 50Hz (referred to 1KHz).

Parts:

P1_________________22K Log.Potentiometer (Dual-gang for stereo)
P2________________100K Log.Potentiometer (Dual-gang for stereo)
R1________________820R 1/4W Resistor
R2,R4,R8____________4K7 1/4W Resistors
R3________________500R 1/2W Trimmer Cermet
R5_________________82K 1/4W Resistor
R6,R7______________47K 1/4W Resistors
R9_________________10R 1/2W Resistor
R10__________________R22 4W Resistor (wirewound)
C1,C8_____________470nF 63V Polyester Capacitor
C2,C5_____________100µF 25V Electrolytic Capacitors
C3,C4_____________470µF 25V Electrolytic Capacitors
C6_________________47pF 63V Ceramic or Polystyrene Capacitor
C7_________________10nF 63V Polyester Capacitor
C9________________100nF 63V Polyester Capacitor
D1______________1N4148 75V 150mA Diode
IC1_____________NE5532 Low noise Dual Op-amp
Q1_______________BC547B 45V 100mA NPN Transistor
Q2_______________BC557B 45V 100mA PNP Transistor
Q3_______________TIP42A 60V 6A PNP Transistor
Q4_______________TIP41A 60V 6A NPN Transistor
J1__________________RCA audio input socket

Power supply parts:

R11_________________1K5 1/4W Resistor
C10,C11__________4700µF 25V Electrolytic Capacitors
D2________________100V 4A Diode bridge
D3________________5mm. Red LED
T1________________220V Primary, 12 + 12V Secondary 24-30VA Mains transformer
PL1_______________Male Mains plug
SW1_______________SPST Mains switch

Notes:

Can be directly connected to CD players, tuners and tape recorders.
Schematic shows left channel only, but C3, C4, IC1 and the power supply are common to both channels.
Numbers in parentheses show IC1 right channel pin connections.
A log type for P2 will ensure a more linear regulation of bass-boost.
Do not exceed 18 + 18V supply.
Q3 and Q4 must be mounted on heatsink.
D1 must be in thermal contact with Q1.
Quiescent current (best measured with an Avo-meter in series with Q3 Emitter) is not critical.
Set the volume control to the minimum and R3 to its minimum resistance.
Power-on the circuit and adjust R3 to read a current drawing of about 20 to 25mA.
Wait about 15 minutes, watch if the current is varying and readjust if necessary.
A correct grounding is very important to eliminate hum and ground loops. Connect to the same point the ground sides of J1, P1, C2, C3 &C4. Connect C9 to the output ground.
Then connect separately the input and output grounds to the power supply ground.

Technical data:

Output power:

10 Watt RMS into 8 Ohm (1KHz sinewave)

Sensitivity:

115 to 180mV input for 10W output (depending on P2 control position)

Frequency response:

See Comments above

Total harmonic distortion @ 1KHz:

0.1W 0.009% 1W 0.004% 10W 0.005%

Total harmonic distortion @ 100Hz:

0.1W 0.009% 1W 0.007% 10W 0.012%

Total harmonic distortion @ 10KHz:

0.1W 0.056% 1W 0.01% 10W 0.018%

Total harmonic distortion @ 100Hz and full boost:

1W 0.015% 10W 0.03%

Max. bass-boost referred to 1KHz:

400Hz = +5dB; 200Hz = +7.3dB; 100Hz = +12dB; 50Hz = +16.4dB; 30Hz = +13.3dB
Unconditionally stable on capacitive loads

Mini-box 2W Amplifier

2009-04-02T01:01:00.000-07:00

Designed for self-powered 8, 4 & 2 Ohm loudspeakers
Bass-boost switch

This amplifier was designed to be self-contained in a small loudspeaker box. It can be feed by Walkman, Mini-Disc, iPod and CD players, computers and similar devices fitted with line or headphone output. Of course, in most cases you will have to make two boxes to obtain stereo.
The circuit was deliberately designed using no ICs and in a rather old-fashioned manner in order to obtain good harmonic distortion behavior and to avoid hard to find components. The amplifier(s) can be conveniently supplied by a 12V wall plug-in adapter. Closing SW1 a bass-boost is provided but, at the same time, volume control must be increased to compensate for power loss at higher frequencies.

In use, R9 should be carefully adjusted to provide minimal audible signal cross-over distortion consistent with minimal measured quiescent current consumption; a good compromise is to set the quiescent current at about 10-15 mA. To measure this current, wire a DC current meter temporarily in series with the collector of Q3.

Parts:

P1_____________10K Log.Potentiometer
R1,R2__________33K 1/4W Resistors
R3_____________33R 1/4W Resistor
R4_____________15K 1/4W Resistor
R5,R6___________1K 1/4W Resistors
R7____________680R 1/4W Resistor
R8____________120R 1/2W Resistor
R9____________100R 1/2W Trimmer Cermet
C1,C2__________10µF 63V Electrolytic Capacitors
C3____________100µF 25V Electrolytic Capacitor
C4,C7_________470µF 25V Electrolytic Capacitors
C5_____________47pF 63V Ceramic Capacitor
C6____________220nF 63V Polyester Capacitor
C8___________1000µF 25V Electrolytic Capacitor
D1___________1N4148 75V 150mA Diode
Q1____________BC560C 45V 100mA PNP Low noise High gain Transistor
Q2____________BC337 45V 800mA NPN Transistor
Q3____________TIP31A 60V 4A NPN Transistor
Q4 ___________TIP32A 60V 4A PNP Transistor
SW1___________SPST switch
SPKR__________3-5 Watt Loudspeaker, 8, 4 or 2 Ohm impedance

Technical data:

Output power:

1.5 Watt RMS into 8 Ohm, 2.5 Watt into 4 Ohm, 3.5 Watt into 2 Ohm (1KHz sinewave)

Sensitivity:

100mV input for 1.5W output @ 8 Ohm

Frequency response:

30Hz to 20KHz -1dB
Total harmonic distortion @ 1KHz & 10KHz:
<0.2%>

3 - 5 Watt Class-A Audio Amplifier

2009-04-02T00:55:00.000-07:00

Behaves like a one-valve operated amplifier
Simple circuitry - No cross-over distortion

In the old valve days, most commercial audio amplifiers suited for compact integrated mono or stereo record players used a one-valve amplifier topology. The circuit was usually implemented by means of a multiple type valve, e.g. a triode pentode ECL86. Common features for those amplifiers were: Class A operation, output power in the 3 - 5W range, input sensitivity of about 600mV for full output power, THD of about 3% @ 3W and 1KHz.

Best types showed THD figures of 1.8% @ 3W and 0.8% @ 2W. This solid-state push-pull single-ended Class A circuit is capable of providing a sound comparable to those valve amplifiers, delivering more output power (6.9W measured across a 8 Ohm loudspeaker cabinet load), less THD, higher input sensitivity and better linearity. Voltage and current required for this circuit are 24V and 700mA respectively, compared to 250V HT rail and 1A @ 6.3V filament heating for valve-operated amplifiers. The only penalty for the transistor operated circuit is the necessity of using a rather large heatsink for Q2 and Q3 (compared to the maximum power delivered). In any case, the amount of heat generated by this circuit can be comparable to that of a one-valve amplifier. An optional bass-boost facility can be added, by means of R5 and C5.

This circuit was built and compared with a one-valve box gramophone circuit of the late 1950s by Aren van Waarde, a Dutch biochemist working in the field of medical imaging (PET) with a strong interest in audio and valve amplifiers. A thorough description of both circuits and the results of subjective test comparisons made by this distinguished Author appeared on AudioXpress magazine: February, March and April 2005 issues.

Parts:

P1_____________47K Log. Potentiometer (Dual-gang for stereo)
R1____________100K 1/4W Resistor
R2_____________12K 1/4W Resistor (See Notes)
R3_____________47K 1/4W Resistor
R4______________8K2 1/4W Resistor
R5______________1K5 1/4W Resistor (Optional, see Notes)
R6______________2K7 1/4W Resistor
R7,R9_________100R 1/4W Resistors
R8____________560R 1/2W Resistor (See Notes)
R10_____________1R 1/2W Resistor
C1,C2__________10µF 63V Electrolytic Capacitors
C3_____________47µF 25V Electrolytic Capacitor
C4____________100µF 35V Electrolytic Capacitor
C5____________150nF 63V Polyester Capacitor (Optional, see Notes)
C6,C7_________220µF 25V Electrolytic Capacitors
C8___________1000µF 25V Electrolytic Capacitor
Q1___________BC560C 45V 100mA Low noise High gain PNP Transistor
Q2,Q3________BD439 60V 4A NPN Transistors
SPKR___________One or more speakers wired in series or in parallel

Total resulting impedance: 8 Ohm
Minimum power handling: 5W

Technical data:

(measured on 8 Ohm resistive load unless otherwise specified)

Sensitivity:

230mV input for 1.5W output
380mV input for 3.5W output
560mV input for 5.6W output

Sensitivity with bass-boost:

400mV input for 1.5W output
630mV input for 3.5W output
850mV input for 5.6W output

Sensitivity with 8 Ohm nominal, loudspeaker cabinet load:

210mV input for 1.5W output
325mV input for 3.5W output
477mV input for 6.9W output

Frequency response:

100Hz to 20KHz 0dB; -3dB @ 40Hz

Frequency response with bass-boost:

+5dB @ 100Hz; +3.9dB @ 200Hz; +2.5dB @ 400Hz; -1dB @ 10KHz and 20KHz

Total harmonic distortion @ 1KHz:

0.3% @ 0.5W; 0.45% @ 1W; 1% @ 5.6W
Unconditionally stable on capacitive loads

Notes:

If necessary, R2 can be adjusted to obtain 13V across C8 positive lead and negative ground.
Total current drawing of the circuit, best measured by inserting the probes of an Avo-meter across the positive output of the power supply and the positive rail input of the amplifier, must be 700mA. Adjust R8 to obtain this value if necessary.
Q2 and Q3 must be mounted on a finned heatsink of 120x50x25mm. minimum dimensions.
Add R5 and C5 if the bass-boost facility is required.

Stereo Preamplifier with Bass-boost

2009-04-01T22:47:00.000-07:00

High Quality, simple design
20 to 30V supply

This preamplifier was designed to cope with CD players, tuners, tape recorders etc., providing an ac voltage gain of 4, in order to drive less sensitive power amplifiers. As modern Hi-Fi home equipment is frequently fitted with small loudspeaker cabinets, the bass frequency range is rather sacrificed. This circuit features also a bass-boost, in order to overcome this problem. You can use a variable resistor to set the bass-boost from 0 to a maximum of +16dB @ 30Hz. If a fixed, maximum boost value is needed, the variable resistor can be omitted and substituted by a switch.

Parts:

P1_________________10K Log.Potentiometer (Dual-gang for stereo)
P2________________100K Log.Potentiometer (Dual-gang for stereo) (See Notes)
R1,R2_____________100K 1/4W Resistors
R3,R6______________15K 1/4W Resistors
R4_________________10K 1/4W Resistor
R5_________________22K 1/4W Resistor
R7__________________1K 1/4W Resistor
R8________________560R 1/4W Resistor
C1,C2,C5____________2µ2 63V Electrolytic Capacitors
C3________________470µF 35V Electrolytic Capacitor
C4__________________1µF 63V Polyester Capacitor
C6_________________47nF 63V Polyester Capacitor
C7_________________22µF 25V Electrolytic Capacitor
IC1_______________TL072 Dual BIFET Op-Amp
SW1________________DPST Switch (Optional, see Notes)

Notes:

Schematic shows left channel only, but R1, R2, R3 and C1, C2, C3 are common to both channels.
For stereo operation P1, P2 (or SW1), R4, R5, R6, R7, R8 and C4, C5, C6, C7 must be doubled.
Numbers in parentheses show IC1 right channel pin connections.
A log type for P2 ensures a more linear regulation of bass-boost.
Needing a simple boost-in boost-out operation, P2 must be omitted and SW1 added as shown in the diagram.
For stereo operation SW1 must be a DPST type.
Please note that, using SW1, the boost is on when the switch is open, and off when the switch is closed.

Technical data @ (30V supply):

Ac voltage gain @ 1KHz: 4
Max. input voltage @ 50Hz: 500mV RMS (280mV RMS @ 20V supply)
Max. input voltage @ 100Hz: 700mV RMS (460mV RMS @ 20V supply)
Max. output voltage: >8V RMS (>5V RMS @ 20V supply)
Max. bass-boost referred to 1KHz:
400Hz = +2dB; 200Hz = +5dB; 100Hz = +10dB; 50Hz = +14dB; 30Hz = +16dB
Total harmonic distortion @ 100Hz and 1V RMS output: 0.02%
Total harmonic distortion @ 1KHz and 1V RMS output: 0.006%
Total harmonic distortion @10KHz and 1V RMS output: 0.007%
Total harmonic distortion @ 100Hz and 5V RMS output: 0.02%
Total harmonic distortion @ 1KHz and 5V RMS output: 0.0013%
Total harmonic distortion @10KHz and 5V RMS output: 0.005%
Current drawing: 2mA

Automatic Loudness Control

2009-04-01T22:42:00.000-07:00

Simple add-on module
Switchable "Control-flat" option

In order to obtain a good audio reproduction at different listening levels, a different tone-controls setting should be necessary to suit the well known behaviour of the human ear. In fact, the human ear sensitivity varies in a non-linear manner through the entire audible frequency band, as shown by Fletcher-Munson curves.

A simple approach to this problem can be done inserting a circuit in the preamplifier stage, capable of varying automatically the frequency response of the entire audio chain in respect to the position of the control knob, in order to keep ideal listening conditions under different listening levels.

Fortunately, the human ear is not too critical, so a rather simple circuit can provide a satisfactory performance through a 40dB range. The circuit is shown with SW1 in the "Control-flat" position, i.e. without the Automatic Loudness Control. In this position the circuit acts as a linear preamplifier stage, with the voltage gain set by means of Trimmer R7. Switching SW1 in the opposite position the circuit becomes an Automatic Loudness Control and its frequency response varies in respect to the position of the control knob by the amount shown in the table below.
C1 boosts the low frequencies and C4 boosts the higher ones. Maximum boost at low frequencies is limited by R2; R5 do the same at high frequencies.

Parts:

P1_________________10K Linear Potentiometer (Dual-gang for stereo)
R1,R6,R8__________100K 1/4W Resistors
R2_________________27K 1/4W Resistor
R3,R5_______________1K 1/4W Resistors
R4__________________1M 1/4W Resistor
R7_________________20K 1/2W Trimmer Cermet
C1________________100nF 63V Polyester Capacitor
C2_________________47nF 63V Polyester Capacitor
C3________________470nF 63V Polyester Capacitor
C4_________________15nF 63V Polyester Capacitor
C5,C9_______________1µF 63V Electrolytic or Polyester Capacitors
C6,C8______________47µF 63V Electrolytic Capacitors
C7________________100pF 63V Ceramic Capacitor
IC1_______________TL072 Dual BIFET Op-Amp
SW1________________DPDT Switch (four poles for stereo)

Technical data:

Frequency response referred to 1KHz and different control knob positions:
Knob position table

Total harmonic distortion at all frequencies and 1V RMS output: <0.01% style="font-weight: bold;">

Notes:

SW1 is shown in "Control flat" position.
Schematic shows left channel only, therefore for stereo operation all parts must be doubled except IC1, C6 and C8.
Numbers in parentheses show IC1 right channel pin connections.
R7 should be set to obtain maximum undistorted output power from the amplifier with a standard music programme source and P1 rotated fully clockwise.

Portable Mixer

2009-04-01T22:29:00.000-07:00

High-quality modular design
9V Battery powered - Very low current drawing

Design description:

The target of this project was the design of a small portable mixer supplied by a 9V PP3 battery, keeping high quality performance. The mixer is formed assembling three main modules that can be varied in number and/or disposition to suit everyone needs.

The three main modules are:

Input Amplifier Module: a low noise circuit equipped with a variable voltage-gain (10 - 100) pre-set, primarily intended as high quality microphone input, also suitable for low-level line input.

Tone Control Module: a three-band (Bass, Middle, Treble) tone control circuit providing unity-gain when its controls are set to flat frequency response. It can be inserted after one or more Input Amplifier Modules and/or after the Main Mixer Amplifiers.

Main Mixer Amplifier Module: a stereo circuit incorporating two virtual-earth mixers and showing the connection of one Main Fader and one Pan-Pot. The image below shows a Block diagram of the entire mixer featuring four Input Amplifier Modules followed by four in-out switchable Tone Control Modules, one stereo Line input, four mono Main Faders, one stereo dual-ganged Main Fader, four Pan-Pots, a stereo Main Mixer Amplifier Module and two further Tone Control Modules switchable in and out for each channel, inserted before the main Left and Right outputs.

Obviously this layout can be rearranged at everyone wish. An astonishing feature of this design lies in the fact that a complete stereo mixer as shown below in the Block diagram draws less than 6mA current!

Mixer Block diagram:

Input Amplifier Module
Parts:

R1,R2,R7_______22K 1/4W Resistors
R3,R4,R5_______47K 1/4W Resistors
R6______________4K7 1/4W Resistor
R8,R13________220R 1/4W Resistors
R9______________2K 1/2W Trimmer Cermet (See Notes)
R10___________470K 1/4W Resistor
R11___________560R 1/4W Resistor
R12___________100K 1/4W Resistor
C1____________470nF 63V Polyester Capacitor
C2,C8_________100µF 25V Electrolytic Capacitors
C3,C4,C5________2µ2 63V Electrolytic Capacitors
C6_____________47pF 63V Ceramic Capacitor
C7______________4µ7 63V Electrolytic Capacitor
Q1____________BC560C 45V 100mA Low noise High gain PNP Transistor
Q2____________BC550C 45V 100mA Low noise High gain NPN Transistor
IC1___________TL061 Low current BIFET Op-Amp

Circuit description:

The basic arrangement of this circuit is derived from the old Quad magnetic pick-up cartridge module. The circuit was rearranged to cope with microphone input and a single-rail low voltage supply.

This low-noise, fully symmetrical, two-transistor head amplifier layout, allows the use of a normal FET input Op-Amp as the second gain stage, even for very sensitive microphone inputs.
The voltage-gain of this amplifier can be varied by means of R9 from 10 to 100, i.e. 20 to 40dB.

Notes:

R9 can be a trimmer, a linear potentiometer or a fixed-value resistor at will.
When voltage-gain is set to 10, the amplifier can cope with 800mV peak-to-peak maximum Line levels.
Current drawing for one Input Amplifier Module is 600µA.
Frequency response is 20Hz to 20KHz - 0.5dB.
Total Harmonic Distortion measured with voltage-gain set to 100: 2V RMS output = <0.02%>
Total Harmonic Distortion measured with voltage-gain set to 10 & 33: 2V RMS output = <0.02%>
THD is much lower @ 1V RMS output.
Maximum undistorted output voltage: 2.8V RMS.

Tone Control

Parts:

P1,P2_________100K Linear Potentiometers
P3____________470K Linear Potentiometer
R1,R2,R3_______12K 1/4W Resistors
R4,R5___________3K9 1/4W Resistors
R6,R7___________1K8 1/4W Resistors
R8,R9__________22K 1/4W Resistors
R10___________560R 1/4W Resistor
R11___________100K 1/4W Resistor
R12___________220R 1/4W Resistor
C1______________1µF 63V Polyester Capacitor
C2_____________47nF 63V Polyester Capacitor
C3,C5___________4n7 63V Polyester Capacitors
C4_____________22nF 63V Polyester Capacitor
C6,C8_________100µF 25V Electrolytic Capacitors
C7______________4µ7 63V Electrolytic Capacitor
IC1___________TL061 Low current BIFET Op-Amp

Circuit description:

This is a straightforward design using the Baxandall-type active circuitry slightly modified to obtain a three-band control. Total voltage gain of this module is 1 when controls are set in their center position.

Notes:

Current drawing for one Tone Control Module is 400µA.
Frequency response is 20Hz to 20KHz - 0.5dB, controls flat.
Tone control frequency range: ±15dB @ 30Hz; ±19dB @ 1KHz; ±16dB @ 10KHz.
Total Harmonic Distortion measured @ 2V RMS output = <0.012%>
THD is below 0.01% @ 1V RMS output.
Maximum undistorted output voltage: 2.5V RMS.

Main Mixer

Parts:

P1,___________100K Linear Potentiometer
P2_____________10K Linear Potentiometer
R1,R2,_________15K 1/4W Resistors
R3,R4,R11,R12_100K 1/4W Resistors
R5,R6__________22K 1/4W Resistors
R7,R8_________390K 1/4W Resistors
R9,R10________560R 1/4W Resistors
R13___________220R 1/4W Resistor
C1,C2_________330nF 63V Polyester Capacitors
C3,C8_________100µF 25V Electrolytic Capacitors
C4,C5__________10pF 63V Ceramic Capacitors
C6,C7___________4µ7 63V Electrolytic Capacitors
IC1___________TL062 Low current BIFET Dual Op-Amp

Circuit description:

The schematic of this circuit is drawn as a stereo unit to better show the input Main Fader and Pan-Pot connections. The TL062 chip contains two TL061 op-amps into the same 8 pin case and is wired as two virtual-earth mixer amplifiers having a voltage gain of about 4, to compensate for losses introduced in the passive Pan-Pot circuitry. Therefore, total voltage-gain is 1. Each channel added to the mixer must include the following additional parts: P1, P2, R1, R2, R3, R4, C1 and C2.

These parts must be wired as shown in the above circuit diagram, connecting R3 and R4 to pin #2 and pin #6 of IC1 for Right and Left channel respectively. These IC1 pins are the "virtual-earth mixing points" and can sum together a great number of channels.

Notes:

Current drawing for one stereo Main Mixer Amplifier Module is 800µA.
Frequency response is 20Hz to 20KHz - 0.5dB.
Total Harmonic Distortion measured @ 2V RMS output = <0.008%>
THD is 0.005% @ 1V RMS output.
Maximum undistorted output voltage: 2.8V RMS.

Further Parts:

To parts listed above should be added: one Main on-off SPST switch, a LED used as pilot-light with its dropping 2K2 1/4W series-resistor, DPDT switches to enable or omit Tone Control Modules as shown in the Block diagram, input and output connectors of the type preferred, one stereo dual-gang 100K potentiometer to fade the Stereo Line Input as shown in the Block diagram, battery clip, PP3 9V battery, knobs etc.

Three-Level Audio Power Indicator

2009-04-01T22:26:00.000-07:00

Battery-operated 3 LED display
Simply connect it to loudspeaker output

Circuit operation:

This circuit is intended to indicate the power output level of any audio amplifier. It is simple, portable, and displays three power levels that can be set to any desired value. For a standard HiFi stereo power amplifier like the 25W MosFet Audio Amplifier described in these pages, the power output values suggested are as follows:

D5 illuminates at 2W
D4 illuminates at 12.5W
D3 illuminates at 24.5W

The above values were chosen for easy setup, but other settings are possible.
IC1A is the input buffer, feeding 3 voltage comparators and LEDs drivers by means of a variable dc voltage obtained by R5 and C4 smoothing action. In order to achieve setting stability, the supply of IC1 and trimmers R6 & R7 is reduced and clamped to 5.1V by Zener diode D1.

Parts:

R1__________100K 1/4W Resistor
R2___________50K 1/2W Trimmer Cermet
R3__________330K 1/4W Resistor
R4____________1M2 1/4W Resistor
R5__________470K 1/4W Resistor
R6,R7_______500K 1/2W Trimmers Cermet
R8____________1K5 1/4W Resistor
R9-R11______470R 1/4W Resistors
C1___________47pF 63V Ceramic Capacitor
C2__________100nF 63V Polyester Capacitor
C3___________47µF 25V Electrolytic Capacitor
C4____________1µF 25V Electrolytic Capacitor
D1______BZX79C5V1 5.1V 500mW Zener Diode
D2_________1N4148 75V 150mA Diode
D3-D5________3mm. Yellow LEDs
IC1_________LM339 Quad Voltage Comparator IC
SW1__________SPST Slider Switch
B1_____________9V PP3
Clip for 9V PP3 Battery

Notes:

The simplest way to connect this circuit to the amplifier output is to use a twisted pair cable terminated with two insulated crocodile clips.
Setup is best accomplished with an oscilloscope or an audio millivoltmeter like the one described in these pages. Precision Audio Millivoltmeter
A 1KHz sine wave generator with variable output is also required (see a suitable circuit in this website also). 1KHz Sinewave Generator
Connect the generator to the amplifier's input and the Audio Power Indicator to the output of the amplifier, in parallel with the oscilloscope probe or the audio millivoltmeter input.
When using high power outputs disconnect the loudspeakers to avoid Tweeters damage and connect in their place an 8 Ohm 20-30 Watt wirewound resistor.
Remember that VRMS output is equal to output Peak-to-Peak Voltage divided by 2.828.
RMS power output in Watts is equal to VRMS2 divided by speaker impedance (usually 8 or 4 Ohm).
Example: set the output of the 1KHz sinewave generator to read 14V on the audio millivoltmeter (24.5W @ 8 Ohm). Set R2 until D3 illuminates, and be sure that D3 turns-off when diminishing a little the generator's output.
Do the same with R7 for D4 and R6 for D5. The readings of the audio millivoltmeter must be 10V (12.5W @ 8 Ohm) and 4V (2W @ 8 Ohm) respectively.

Six-LED Bar Power Indicator

2009-04-01T22:21:00.000-07:00

Useful to monitor audio power delivered to loudspeakers
No power supply - no setup required

This device, connected to the loudspeaker output of an audio amplifier, will indicate the instantaneous output power delivered to the loudspeaker(s) by means of six LEDs illuminating one after another by voltage values increasing little by little, providing the visual impression of a luminous bar or column, increasing and decreasing in height following the increase and decrease of the signal's level.

The input signal is first rectified by D1 and then sent to six different voltage dividers, one for each LED. In this way, the indication provided by the LEDs illumination of this "Power Display", will be related to the instantaneous power sunk by the whole loudspeaker cabinet. Six output power levels are displayed by the LEDs in a 2W - 80W range (no setup required). Each nominal power level indication into 8 Ohms load is reached when the respective LED illuminates at full brightness.

Parts:

R1_____________220R 1/2W Resistor
R2,R5,R6,R8____100R 1/4W Resistors
R10,R12,R14____100R 1/4W Resistors
R3_____________220R 1/4W Resistor
R4,R7__________330R 1/2W Resistors
R9_____________560R 1/2W Resistor
R11____________820R 1/2W Resistor
R13______________1K2 1/2W Resistor
D1___________1N4004 400V 1A Diode
D2,D4,D6__BZX79C2V7 2.7V 500mW Zener Diodes
D3,D5,D7,D8,D9,D10 Red LEDs (Any dimension and shape) (See Notes)

Notes:

The output power indicated by each LED must be doubled when 4 Ohms loads are driven.
The circuit can be adapted to suit less powerful amplifiers by reducing the number of LEDs and related voltage dividers.
LEDs of any dimension can be used, but rectangular shaped devices will be more suitable to be compacted in bars or columns.
For a stereo amplifier, two identical circuits are required.

Precision Audio Millivoltmeter

2009-04-01T20:34:00.000-07:00

Measures 10mV to 50Volt RMS in eight ranges
Simply connect to your Avo-meter set to 50µA range

Parts:

R1_____909K 1/2W 1% Metal Oxide Resistor
R2______90K9 1/2W 1% Metal Oxide Resistor
R3_______9K09 1/2W 1% Metal Oxide Resistor
R4_______1K01 1/2W 1% Metal Oxide Resistor
R5_____100K 1/4W Resistor
R6_______2M2 1/4W Resistor
R7______82K 1/4W Resistor
R8______12K 1/4W Resistor
R9_______1K2 1/4W Resistor
R10______3K3 1/4W Resistor
R11____200R 1/2W Trimmer Cermet
C1_____330nF 63V Polyester Capacitor
C2,C3__100µF 25V Electrolytic Capacitor
C4_____220µF 25V Electrolytic Capacitor
C5______33pF 63V Polystyrene Capacitor
C6_______2µ2 63V Electrolytic Capacitor
D1-D4___1N4148 75V 150mA Diodes
IC1_____CA3140 Op-amp
IC2_____CA3130 Op-amp
SW1_____2 poles 5 ways rotary switch
SW2_____SPDT switch
J1______RCA audio input socket
J2,J3___4mm. output sockets
B1______9V PP3 Battery
Clip for PP3 Battery

Notes:

Connect J2 and J3 to an Avo-meter set to 50µA range:
Switching SW2 the four input ranges will be multiplied by 5
Total fsd ranges are: 10mV, 50mV, 100mV, 500mV, 1V, 5V, 10V, 50V
Set R11 to read 1V in the 1V range, with a sine wave input of 1V @ 1KHz
Compare the reading with that of another known precision Millivoltmeter or with an oscilloscope.
The oscilloscope reading must be a sinewave of 2.828V peak to peak amplitude
Frequency response is flat in the 20Hz-20KHz range
If you have difficulties in finding resistor values for R1, R2, R3 & R4, you can use the following trick:

R1 = 10M + 1M in parallel
R2 = 1M + 100K in parallel
R3 = 100K + 10K in parallel
R4 = 1K2 + 6K8 in parallel
All resistors 1/4W 1% tolerance

Sound Pressure Level Meter

2009-04-01T20:30:00.000-07:00

Suitable for home-cinema setup
9 Volt operated portable unit

This project was requested by some correspondents wishing to setup their home-cinema set adjusting all the loudspeaker outputs to the same level when heard from the listening position.
In practice this device is a simple (though linear and precise) ac millivoltmeter, using an existing multimeter set to 50 or 100µA fsd with the probes connected to J1 and J2 to read the results.
The precision of the measure is entirely depending on the frequency response of the microphone used but, fortunately, for the main purpose of this circuit an absolutely flat response is not required. Therefore, a cheap miniature electret microphone can be used.

Use:

The amplifiers driving the loudspeakers must be fed, one at a time, with a sine wave in the 400Hz - 1KHz range, but different values can also be chosen. For this purpose you can use a simple signal generator circuit like one of those available on this site, namely: 1KHz Sine wave Generator or, better still, Spot-frequency Sine wave Generator. As an alternative, the input sine wave can be provided by a CD test track, a cassette-tape or a personal computer.

Please be careful and set the volume control very low, to avoid loudspeakers' damage. Switch-on the Sound Pressure Level Meter and increase the volume of the amplifier in order to obtain an approximate center-scale reading. Repeat the same steps with all channels.

Parts:

R1______________10K 1/4W Resistor
R2,R3___________22K 1/4W Resistors
R4_____________100K 1/4W Resistor
R5_____________100R 1/4W Resistor
C1_______________1µF 63V Polyester or Electrolytic Capacitor
C2_____________100µF 25V Electrolytic Capacitor
C3_____________220µF 25V Electrolytic Capacitor
D1-D4_________BAT46 100V 150mA Schottky-barrier Diodes
IC1__________CA3140 Op-Amp IC
MIC___________Miniature electret microphone (See Notes)
J1,J2___________4mm Output sockets
SW1____________SPST Toggle or Slider Switch
B1______________9V PP3 Battery
Clip for PP3 Battery

Notes:

If external electret (battery powered) or dynamic microphone types are used, R1 must be omitted.
Using a good quality microphone, this circuit can be very useful in setting-up parametric equalizers or tone controls in HiFi chains etc.

1KHz Sine wave Generator

2009-04-01T20:27:00.000-07:00

Simple circuitry, low distortion, battery operated
Variable, low impedance output up to 1V R

Circuit description:

This circuit generates a good 1KHz sinewave adopting the inverted Wien bridge configuration (C1-R3 & C2-R4). It features a variable output, low distortion and low output impedance in order to obtain good overload capability. A small filament bulb ensures a stable long term output amplitude waveform. Useful to test the Precision Audio Millivoltmeter, Three-Level Audio Power Indicator and other audio circuits posted to this website.

Parts:

R1____________5K6 1/4W Resistor
R2____________1K8 1/4W Resistor
R3,R4________15K 1/4W Resistors
R5__________500R 1/2W Trimmer Cermet
R6__________330R 1/4W Resistor
R7__________470R Linear Potentiometer
C1,C2________10nF 63V Polyester Capacitors
C3__________100µF 25V Electrolytic Capacitor
C4__________470nF 63V Polyester Capacitor
Q1,Q2_______BC238 25V 100mA NPN Transistors
LP1___________12V 40mA Filament Lamp Bulb (See Notes)
J1__________Phono chassis Socket
SW1__________SPST Slider Switch
B1_____________9V PP3
Clip for 9V PP3 Battery

Notes:

The bulb must be a low current type (12V 40-50mA or 6V 50mA) in order to obtain good long term stability and low distortion.
Distortion @ 1V RMS output is 0.15% using a 12V 40mA bulb, raising to 0.5% with a 12V 100mA one.
Using a bulb differing from specifications may require a change of R6 value to 220 or 150 Ohms to ensure proper circuit's oscillation.
Set R5 to read 1V RMS on an Audio Millivoltmeter connected to the output with R7 rotated fully clockwise, or to view a sinewave of 2.828V Peak-to-Peak amplitude on the oscilloscope.
With C1, C2 = 100nF the frequency generated is 100Hz and with C1, C2 = 1nF frequency is 10KHz but R5 requires adjustment.
High gain transistors are preferred for better performance.

Low-distortion Audio-range Oscillator

2009-04-01T20:23:00.000-07:00

Generates very low-distortion sine waves up to 1V RMS
No thermistors required - No settling time

Producing low-distortion sine waves, this oscillator operates over the range 16 to 22000 Hz.
The circuit is based on two articles that have appeared earlier in Wireless World - Roger Rosens' "Phase -Shifting Oscillator", February 1982 pp. 38-41, and J. L. Linsley Hood's "Wien-Bridge Oscillator with low harmonic distortion" from May 1981 pp. 51-53.

This design features the simplicity of the Rosens' circuit but avoids the use of a thermistor. Instead, oscillator stability is controlled by means of a common photo-resistor driven by a LED, as suggested in the Linsley Hood article. There is no settling time when the oscillator's frequency is changed and no bouncing of the output waveform. Use of an expensive and sometimes difficult to obtain thermistor is avoided.

Parts:

P1_____________10K Log. Potentiometer (Dual-ganged)
P2______________2K2 Linear Potentiometer
R1,R2,R4,R5_____3K3 1/4W Resistors
R3,R6_________820R 1/4W Resistors
R7_____________10K 1/2W Trimmer Cermet
R8_____________22K 1/4W Resistor
R9_____________Photo resistor (any type)
R10_____________8K2 1/4W Resistor
R11,R12,R14,R15_3K3 1/4W Resistors
R13_____________2K7 1/4W Resistor
R16--R20________3K3 1/4W Resistors
R21____________56K 1/4W Resistor
R22____________68K 1/4W Resistor
R23_____________1K 1/4W Resistor
C1,C6_________220pF 63V Polystyrene Capacitors
C2,C7___________8n2 63V Polyester Capacitors
C3,C8__________82nF 63V Polyester Capacitors
C4,C9_________150nF 63V Polyester Capacitors
C5,C10________680nF 63V Polyester Capacitors
D1--D4______1N4148 75V 150mA Diodes
D5_____________LED 5mm. Red
IC1,IC2_____NE5532 Low noise Dual Op-amps
IC3__________TL084 Quad BIFET Op-Amp
SW1__________2 poles 3 ways rotary switch

Technical data:

Output voltage: Sine wave, 1V RMS max.
Total harmonic distortion @ 1V RMS output:

Frequency Reading
100Hz = 0.0035%
300Hz = 0.0028%
1kHz = 0.002 %
3kHz = 0.002 %
10kHz = 0.001 %

Notes:

Any common photo-resistor and 5mm. red LED can be used, provided they are in close contact and enclosed in a light-proof small box. I used the metal screen of a small IF transformer for AM transistor radios sealed with black insulating tape.
The 10K trimmer must be set to obtain a 1V RMS output.
The circuit must be supplied by a + and - 15V dual regulated supply. Common 7815 and 7915 regulator ICs should be used for this purpose.

Simple Square wave Generator

2009-04-01T20:20:00.000-07:00

Three switchable frequencies: 100Hz, 1KHz, 10KHz
1.5V battery operated, minimum parts counting

This simple circuit generates a good and stable 1V peak-to-peak square wave at 100Hz, 1KHz and 10KHz using a single 1.5V cell as power supply.

An useful feature of this circuit is that frequency changes can be obtained by switching only one capacitor at a time. Current consumption is about 600µA.

Parts:

R1____________560K 1/4W Resistor
R2____________680R 1/4W Resistor
R3______________2K2 1/4W Resistor
R4____________150K 1/4W Resistor
C1_____________12nF 63V Polyester Capacitor
C2______________1n2 63V Polyester Capacitor
C3____________120pF 63V Polystyrene or ceramic Capacitor
C4,C5__________10µF 25V Electrolytic Capacitors
Q1,Q2________BC549C 25V 100mA NPN High-gain Low-noise Transistors
SW1____________SPST Slider Switch
SW2____________1 pole 3 ways Rotary Switch
B1_____________1.5V Battery (AA or AAA cell etc.)

Notes:

If a precise 50% duty-cycle is needed, trim R1 and monitor the output wave form by means of an oscilloscope.
A good 500mV peak-to-peak square wave is provided even at 1V supply.

Self-powered Sine to Square wave Converter

2009-04-01T20:14:00.000-07:00

Converts sine to square waves without a power-source
Useful as a test instrument for audio purposes

This circuit is intended to provide good square waves converting a sine wave picked-up from an existing generator. Its main feature consists in the fact that no power-source is needed: thus it can be simply connected between a sine wave generator and the device under test.

The input sine wave feeds a voltage doubler formed by C1, C2, D1 & D2 that powers the IC. IC1A amplifies the input sine wave, other inverters included in IC1 squaring the signal and delivering an output square wave of equal mark/space ratio and good rise and fall times through the entire 20Hz-20KHz range.

Parts:

R1_____________1M 1/4W Resistor
R2___________100K Linear Potentiometer
C1,C2________100µF 25V Electrolytic Capacitors
C3____________10nF 63V Polyester Capacitor
D1,D2_______1N4148 75V 150mA Diodes
IC1___________4069 Hex Inverter IC

Notes:

Best performances are obtained with an input sine wave amplitude from 1V RMS onwards.
Output square wave amplitude is proportional to input amplitude.
Minimum sine wave input amplitude needed for good performance: 750mV RMS.
Output square wave amplitude with 1V RMS input: 3V peak to peak, with R2 set at max.
Minimum output square wave amplitude: 2V peak to peak, with R2 set at max.
Substituting the two silicon diodes with germanium types (e.g. AA118, AA119), the minimum input threshold can be lowered.